Abstract commonly expected that very distortion amplifiers must dissip

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Abstract commonly expected that very distortion amplifiers must dissip

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Pushing Quiescent Power Amps Greater than 55dBM 2-Tone Intercept
Abstract commonly expected that very distortion amplifiers must dissipate considerable quiescent power achieve their high linearity. With advent intrinsically distortion current feedback amps, along with linearity improvements achieved negative feedback used these devices, wideband distortion amplifiers have become available much lower quiescent power levels. This discussion will focus 2-tone, order intermodulation distortion current feedback amplifiers. Following brief review harmonic distortion mechanisms current feedback amps, simple means further improve already high intercept will described. Having pushed order spurious levels into noise, automated means measuring these very distortion levels will described.
Input Transimpedance Gain Stage Buffer Output Buffer
code error current signal through their collectors current mirror stage. outputs symmetric current mirror stages back together form high transimpedance node amplifier. This high gain node amplifier. Small changes error current (fed back through will have significant transimpedance gain voltage outputs current mirrors. This voltage (Vo') buffered output another open loop voltage buffer, transistors Q5-Q8. This buffer's high input impedance contributes achieving high forward transimpedance gain through amplifier while providing impedance output drive. Both input buffer output buffer essentially Class buffer stages (see ref. more complete description current feedback amp). this point, amplifier's internal elements have been treated from open loop standpoint. When output connected back inverting input through feedback resistor, with gain setting resistor ground inverting node, closed loop configuration. Figure shows closed loop current feedback block diagram along with resulting transfer function.
+Vcc
Current Mirror
-Vcc +Vcc ierr Current Mirror -Vcc +Vcc
Z(s)ierr
ierr
Where
Z(s)
-Vcc
desired inverting gain
Z(s)
Figure Simplified current feedback topology Harmonic Distortion Current Feedback Amplifier Figure shows simplified internal circuit current feedback operational amplifier. Note that structure very symmetric with complementary devices. input buffer stage, Q1-Q4, forms open loop voltage buffer from non-inverting input inverting pin. Transistors provide means simultaneously drive inverting node voltage cas© 1993 National Semiconductor Corporation
Printed U.S.A.
frequency dependent loop gain
Figure Closed-loop transfer function Looking transfer function, numerator expression desired signal gain Volts/Volts. While denominator expression represents error terms finite forward gain amplifier. forward transimpedance gain, Z(s), were infinite, this error term
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would drop amplifier would produce exactly gain shown numerator. forward transimpedance however, frequency dependent gain, having very large value with dominant frequency pole along with higher frequency poles (see National op-amp data sheets open loop transimpedance plots). When magnitude Z(s) rolled equal value feedback transimpedance Rf/Rg)), loop gain dropped overall amplifier frequency response begins roll off. Rf/Rg) part feedback transimpedance principal parasitic effect limiting amplifier's bandwidth higher closed loop Rf/Rg) gains desired output impedance buffer driving inverting node). remainder this discussion this term will zero leaving feedback transimpedance only basic advantages offered current feedback topology that, with loop gain, hence frequency response, externally desired signal gain with minimal impact frequency response. through this mechanism that current feedback said offer GainBandwidth product "independence". Please National application note OA-14 more complete discussion current feedback transfer function frequency response control. Distortion Mechanism's Current Feedback Amplifiers From open loop standpoint, harmonic distortion arises from non-linearities going from inverting error current signal output voltage. Although have shown this transimpedance gain frequency dependent linear gain, Z(s), particular frequency, actually represented polynomial expression from error current output voltage. This polynomial will have very high linear gain term frequencies) with relatively small coefficients higher order terms. signal path from inverting input output follows symmetric paths shown Figure open loop order coefficient transfer mismatches between upper lower signal paths output. open loop order coefficient principally order curvature (crossover distortion) transfer function Class output buffer. (See reference page discussion Class buffer distortion). With order distortion arising from mismatch effects, often observed that this distortion strongly dependent operating point output. Changing relative voltages across halves forward gain path will effect balance parasitic effects (voltage dependent base-collector capacitance output impedances) that give rise this non-linearity. Similarly, ground centered, sinusoidal, output swing, inbalancing power supplies used null harmonic specific frequency.
magnitude open loop harmonic distortion term, given frequency, principally function output load current biasing current, Ib2, output stage. Hence, signal swings load resistors down, quiescent biasing current goes down, this third order distortion will increase. Conversely, load impedance increases, signal swing decreases, quiescent biasing current increases, this order distortion will decreased. intrinsic symmetry forward gain path, with well matched signal paths, along with fully complementary Class output buffer, yields open loop distortion. This distortion further reduced action negative feedback when loop closed shown Figure particular frequency, Z(s) taken have specific values coefficients polynomial approximation transimpedance gain from inverting error current output voltage. Figure steps through transfer function development using polynomial expression forward transimpedance gain. Although this approach does yield closed form solution output voltage polynomial input signal, does illustrate loop gain dependence higher order terms.
f(ierr
linear transimpedance order gain order gain
currents inverting node ierr From above ression first three terms) ierr Should isolate solve ierr polynomial here, this doesn' yield very clear result. Simply putting this ierr ression above ression multiplying through grouping terms
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Then, solving where Loop gain
shows Bode plot frequency dependent forward transimpedance, 20*log(|Z(s)|), typical current feedback amplifier. solid horizontal line intersecting this plot about 100MHz 20*log(Rf), feedback transimpedance. vertical distance between forward Z(s) this solid horizontal line loop gain, |Z(s)|/Rf.
Figure Loop gain effect non-linearity more common show effect negative feedback forward gain distortion effects introduce error signal output forward gain block. Figure shows this control theory approach with similar result Figure open loop distortion effects reduced loop gain negative feedback closed loop configuration. This approach also does reach closed form solution should actually depend Vi).
Differenting Stage Feedback Ratio Verr Distortion Signal Forward Gain
Z(s) Transimpedance (dBW) (10dB/div) Loop Gain Unity loop gain crossover
(Rf)
Frequency (Hz)
Figure Forward feedback transimpedance
Verr Verr
Loop Gain Figure Control theory model distortion single frequency input, loop gain fundamental frequency that applicable determining loop gain's impact distortion. Some literature seems imply that loop gain harmonics that acting linearize closed loop performance decrease distortion (reference page 418). However, testing with loop gain tailored higher fundamental than harmonics shown direct dependence loop gain fundamental, harmonics. order terms, achievable harmonic 2-tone, intermod, distortion levels intrinsic order distortion output buffer loop gain fundamental frequency operation. Figure
apparent from Figure this loop gain decreases with increasing frequency forward transimpedance gain rolls off. intersection Z(s) feedback transimpedance critical importance determining closed loop frequency response flatness. This decreasing loop gain with frequency dominant cause increase harmonic distortion with increasing frequency negative feedback amplifiers. contributions current feedback amplifier ability higher loop gains higher frequencies than equivalent voltage feedback parts. Typically, however, still order distortion terms starting increase, given output power level gain, frequencies above 5MHz. Conversely, operating frequencies below about 5MHz, distortion performance typically reaches minimum value region dominant open loop pole. point comparison, this discussion will focus 2-tone order intermodulation distortion 10MHz. Figure shows plot order intercept 10MHz) quiescent power dissipation dBm) current feedback amplifiers along with several other high linearity amplifiers. Many these other amplifiers Class output which requires significantly higher quiescent power achieve distortion. Also, minimal feedback, hence minimal distortion improvement loop gain, generally used these other parts. This yields distortion performance that nearly frequency dependent. Basically, these Class output amplifiers have driven forward path non-linearities down with high quiescent currents used minimal feedback keep their distortion performance constant over wider frequency range.
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Order Intercept (dBm)
relatively large inverting input current noise current feedback amplifier dominate overall noise performance. This noise current shows output multiplied feedback resistor. circuit Figure reduces feedback resistor using parallel combination feedback resistors frequency gain this noise current. coupling inductor remove this parallel feedback path before unity loop gain crossover frequency (where Z(s) Rf1) with approximately degree phase margin.
Current Feedback
Quiescent Power Dissipation (dBm)
Recommended feeback resistor value specific
National National Advancd Milliwave Adams Russell Avantek Watkins Johnson Q-Bit
CLC221 CLC401 ARO1003252 AM-109 UTO-509 WJ-A59 QB-210
Figure Loop gain shaping network Figure shows generalized transfer function circuit Figure general, this circuit could used shape both loop gain forward signal gain. frequencies, gain parallel combination Rf's divided parallel combination Rg's. high frequencies, once inductor opened connection between feedback's, gain simply Rf1/Rg1). Similarly, feedback transimpedance frequencies parallel combination feedback while high frequencies increased equal Rf1. going from high frequencies, this circuit shows zero/pole pair both signal gain feedback transimpedance.
Figure 2-tone, order intercept quiescent power (dBm) data Figure shows that, frequencies, considerably higher intercepts quiescent dissipation achieved with current feedback amps through very linear forward gain path high loop gain feedback network. principal drawbacks using current feedback application steadily decreasing intercept above 5MHz, usable bandwidths limited about 100MHz, relatively poor noise performance. Intercepts have typically dropped below 30dBm 50MHz National amps shown Figure (note Improving Distortion Shaping Loop Gain Since current feedback amplifier allows loop gain separately from signal gain, should possible adjust loop gain yield improved distortion performance without changing signal gain. simplest this would scale resistor values down, keeping same ratio Rf/Rg. Decreasing will increase loop gain over full frequency range, runs risk inadequate phase margin crossover, where |Z(s)| would preferable decrease feedback transimpedance lower frequencies return nominal design value where this feedback intended equal Z(s). Figure shows possible circuit that achieves this loop gain shaping. This circuit originally reported literature means improve equivalent input noise (reference closed loop gain settings,
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Figure Transfer function parallel inductor coupled feedback
this loop gain shaping application, will take ratio Rf1/Rg1 Rf2/Rg2. Under this condition, numerator transfer function Figure simplifies equal Rf1/Rg1 indicating flat frequency response roll-off frequency. principle concern here frequency dependence feedback transimpedance. loop gain been increased decreased feedback transimpedance frequencies which should provide improved harmonic distortion performance. feedback transimpedance shows zero/pole pair inductor coupling feedback paths together. Figure shows analysis loop gain this parallel, inductor coupled, feedback pole zero frequencies.
With
pole frequency feedback transimpedance less than nominal unity loop gain crossover frequency selected amp. desired reduction feedback transimpedance frequencies set. This will also determine zero frequency. pole/zero ratio equal
Normally
coupling inductor solved from equation Figure solved using desired signal gain. additional loading output additional feedback network (Rf2 Rg2) should checked that significantly lowering intended load. Figure shows Bode plot same forward transimpedance gain Figure with reduction feedback transimpedance using this paralleled, inductor coupled, feedback. Note that targeted pole frequency 80MHz which forces zero frequency 20MHz. This yields 12dB times) increase frequency loop gain. This loop gain decreased 20MHz continues only improvement from nominal value 80MHz. goal here approximately back feedback impedance 100MHz unity loop gain crossover point forward transimpedance curve. This 12dB improvement loop gain 10MHz should translate directly into increase 2-tone, order intermodulation intercept (one half 12dB decrease spurious levels given output power level will yield increase intercept).
Z(s)
Rewriting Loop Gain terms Loop gain Z(s) forward transimpedance feedback transimpedance
high frequencies, feedback transimpedance Define ratio this high frequency feedback transimpedance feedback transimpedance
then increase frequency loop gain from high frequency value resulting feedback transimpedance zero pole ratio
Typically would target pole frequency occur lower frequency than nominal crossover frequency (recommended value amplifier) Then Pole frequency This will zero frequency pole frequency
Gain
Loop Gain
Frequency (MHz)
Figure Loop gain analysis parallel inductor coupled feedback this approach increasing frequency loop gain following steps would followed: desired signal gain would set. current feedback amplifier appropriate this gain range would selected. selected amp's nominal recommended value (note
Figure Forward transimpedance with shaped feedback showing increased loop gain Test Circuit Implementing Enhanced Frequency Intercept Figure shows example circuit using paralleled feedback approach increasing loop gain frequencies. This circuit also uses step transformer input improve Noise Figure (reference
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This input stage presents input impedance passband transformer reduces overall Noise Figure this test circuit. Although this transformer will couple signal path, important remember that amplifier itself true coupled device. Since input already coupled, blocking capacitor output been added strip amplifier offsets that present. itself presents very output impedance. into system, series, discrete, resistor must added. defined measurement point both gain intercept load.
+15V 6.8µF Input Impedance 3.9pF 6.8µF -15V 1500 200nH 78.9 26.1 (internal) 20V/V 0.1µF 0.01µF
900mW quiescent power. Using volt supplies, full scale output voltage swing approximately volts order satisfy 50mA maximum output current into load. 2-tone, order intercept testing, this translates into maximum 12dBm test power level each test frequencies load. Given maximum peak peak swing amplifier output, from either voltage swing current limit standpoint, maximum single tone power level load voltage swing this level. This accounts loss going through matching network fact that full voltage envelope tone test peak peak swings individual test frequencies. Pushing full 12dBm each test tone load puts amplifier into slightly higher distortion mode. 49dBm intercept amplifier itself observed until single tone power load dropped 8dBm. Using this test condition will yield 3Vpp swing voltage envelope load. This test circuit shows bipolar supplies amplifier. True coupled devices typically balanced bipolar supplies. However, since most current feedback amplifiers don't actually ground reference their design, single supply operation perfectly acceptable. National application note OA-11 describes this detail. Measuring 2-Tone, Order Intercepts above 50dBm order spurious levels 8dBm test power levels 50dBm intercept would -74dBm (note This 82dB dynamic range requirement right edge what most spectrum analyzers measure. Figure shows typical test setup measuring intercepts where dynamic range requirement less than 90dB. principle requirements this measurement provide clean input test tones, with intermodulation sources. amplifiers following sources hybrid power combiner achieve this quite well. These amplifiers, CLC142, provide 27dBm output power through 100MHz particularly useful getting enough power testing gain devices. They also isolate leveling loops output stage sources from mixing together themselves provide excess 50dBm intercepts through 10MHz (although their intercept performance does limit this test). primary requirement test setup Figure attenuate fundamental power levels input spectrum analyzer mixer enough power level eliminate analyzer produced intermodulation products. This typically translates into -40dBm power level mixer.
CLC221
load
0.01µF
0.1µF
Figure Test circuit increasing loop gain overall gain this circuit 40V/V (32dB). Figure shows measured frequency response this test circuit. Although CLC221 specified provide >165MHz -3dB bandwidth gain configuration used here, transformer input filter limit bandwidth approximately 90MHz. Significant latitude trading gain, bandwidth, noise possible when using amps these types applications. Please National application note OA-11 additional discussion interpreting using specifications applications.
Magnitude (1dB/div)
Gain 31.75dB 38.7V/V -3dB 91MHz
Frequency (10MHz/div)
Figure Measured frequency response wide dynamic range test circuit using CLC221 10MHz, CLC221, without loop gain shaping, 49dBm order intercept while dissipating only
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test Figure would begin have trouble making this measurement required dynamic range extends beyond 85dB. This would occur test power level decreased evaluate intercept performance observed) higher intercepts were measured. anticipated 55dBm intercept test circuit described earlier would exceed measurement range
Frequency (100KHz 10MHz) Order Intermodulation Intercept Test Setup Template
15dB gain
Fluke 6080A
CLC142 Gain
-3dB IMD3 Computation -3dB
Gain
Hybrid Power Combiner
-35dB
HP8568B Spectrum Analyzer Internal Atten. 10dB
Mini-Circuits ZFSC-2-6
ATTEN
Gain Fluke 6080A CLC142
15dB gain
Into Spectrum -28dBm Analyzer
Figure Intermod test setup
Frequency (100KHz 10MHz) Order Intermodulation Intercept Test Setup Template With Output Single Tone Cancellation
Fluke 6080A
CLC142
Gain
Gain
Hybrid Power Combiner Mini-Circuits ZFSC-2-6
-3dB -35dB -3dB Hybrid Power Combiner Gain Mini-Circuits ZFSC-2-6
IMD3 Computation HP8568B Spectrum Analyzer Internal Atten. 10dB
-3dB -35dB -3dB 10dB
Fluke 6080A
CLC142
Fluke 6080A
CLC142
Phase Locked
Figure Improved dynamic range intermod test setup test configuration Figure With 8dBm test power levels load, anticipated spurious power levels would -86dBm 55dBm intercept. This 94dB dynamic range probably beyond most spectrum analyzers. Averaging could used lower noise floor attempt pull this very spurious noise. test power decreased, required averaging would significantly extend test time. alternative technique would filter both test frequencies after before signal applied analyzer. This realistically requires relatively broad spacing test input frequencies avoid filtering spurious frequencies) particularly useful swept frequency measurements. Figure shows modification basic test setup Figure extend dynamic range this intermodu7 lation measurement system. this approach signal source, phase locked other input signal sources, used null test frequency output DUT. same type power combiner used input used here combine output signal with nulling signal from signal generator. Fluke 6080A sources offer programmable phase capability that allows nulling source tuned very nearly degrees phase with test signals output. output this power combiner will then have test frequencies plus original intermodulation tones output DUT. This allows system less attenuation analyzer mixer since intermodulation terms will generated analyzer. This technique also offers advantage being fully programmable over wide range frequencies test powers.
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Test Results Improved Intercept Amplifier intermodulation intercept circuit Figure first measured using basic setup Figure This measurement resulted approximately 52dBm intercept. Figure shows test signals ±100kHz around 10MHz. Note that measured power -27dBm -37dBm mixer considering analyzer's internal 10dB attenuator.
0.0dBm 10dB/ CENTER 10.0MHz ATTEN 10dB 9.0MHz -27.0dBm
absolute power level upper tone, -21dBm, since lower tone significantly lower, analyzer generated intermodulation terms should interfere with this measurement.
9.90MHz -51.90dBm CENTER 10.0MHz
0.0dBm 10dB/
ATTEN 10dB
Power (dBm)
Power (dBm)
CENTER 10.0MHz 10MHz
10Hz
SPAN 1.0MHz 200sec
CENTER 10.30MHz 10MHz
10Hz
SPAN 1.0MHz 200msec
Figure Test power levels analyzer with single-tone cancellation Figure shows measured lower spurious 9.7MHz using test setup Figure This measured spurious level about more noise than earlier measurement. However, noise floor clearly come along with this reduced attenuation from output analyzer input. This indicates that output noise significant part total noise analyzer. Taking measured test tone power 11dBm, while output power load 8dBm, this yields (-11-(-98.5)) 51.75dBm intercept.
9.70009MHz -98.50dBm LEVEL -40.0dBm
Figure Test power levels 10MHz test Figure shows measurement upper spurious signal 10.3MHz. Note very narrow resolution bandwidth, video bandwidth, span make this measurement. phase noise sources phase locking sources analyzer together critical this measurement.
-40.0dBm 10dB/ CENTER 10.30MHz ATTEN 10dB 10.300010MHz -113.10dBm
-40.0dBm 10dB/
ATTEN 10dB
Power (dBm)
CENTER 10.30MHz 10MHz
10Hz
SPAN 300MHz 10sec CENTER 9.70MHz 10MHz 10Hz SPAN 300MHz 10sec
Figure Spurious measurement 10.3MHz Continuing same measurement with test Figure allow slightly increased measurement range. After nulling lower test frequency, test signal plot Figure shows widely disparate power levels going into analyzer. important remember that equal test power levels still being generated output. Note that measured power un-cancelled test tone increased approximately 11dBm from earlier -27dBm level. This reflects reduced (19dB) attenuation used output signal path. Also note that lower test tone been attenuated 40dB from upper tone with this cancelling technique. Although mixer seeing fairly high
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Figure Spurious measurement with single test tone cancellation loop gain shaping network Figure appears have improved intercept only 3dBm instead 6dBm expected. Further investigation revealed that this attributed non-zero inverting input impedance current feedback amplifier. With this impedance some feedback error signal splits wasted through gain setting resistors. This reduces loop gain. Simulations including this effect revealed only improvement loop gain 10MHz which consistent with 3dBm increase intermodulation intercept.
Power (dBm)
However, intercept does continue improve test frequency decreased. following tables shows measured intercepts from 5MHz 10MHz circuit Figure using test setup Figure Frequency Intercept 5MHz 6MHz 7MHz 8MHz 9MHz 10MHz 55.8dBm 54.9dBm 54.0dBm 52.8dBm 52.4dBm 51.5dBm
References: "Current Feedback Amplifiers", Sergio Franco; reprinted National 1993 Databook Application Note Integrated Electronics: Analog Digital Circuits Systems, Millman Halkias; McGraw/Hill 1972 Network Quiets Current-Feedback Amps", Howard Bandell; EDN, Aug. 1990 page "Improving Amplifier Noise Figure High Order Intercept Amplifiers", National Semiconductor Application Note OA-14 Notes: 49dBm intercept shown CLC221 does agree with data sheet plot showing 55dBm 10MHz. data Figure taking signal power load through series output matching resistor. CLC221 data sheet plot taking test power output which yields 6dBm higher intercept. More recent data sheets, this discussion, taking signal power matched load. National Semiconductor Application note OA-13 discussion setting various signal gains. Equal test powers spurious test Intercept dBm. 8dBm test 49dBm intercept, spurious
Below 5MHz 56dBm intercept achieved. Conclusions: High speed current feedback amplifiers offer exceptional 2-tone order intercept performance relatively quiescent powers. This intercept performance does decrease with frequency decreasing loop gain frequency increased. Although example device used here, CLC221, high performance hybrid amplifier, similar results lower maximum output power levels achieved with monolithic current feedback amplifiers (such CLC409, CLC401 CLC404 particularly). simple loop gain shaping network used further increase intercept frequencies. finally, simple means extend measurement dynamic range through output test signal cancellation been described demonstrated. This approach offers principle advantage being easily programmable over wide range frequencies.
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Abstract commonly expected that very distortion amplifiers must dissip

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