HV91 Series Application Note AN-H13 Designing High-Performance Flyback
|
|
|
Top Searches for this datasheet
HV91 Applications
HV91 Series Application Note AN-H13 Designing High-Performance Flyback Converters with HV9110 HV9120
Ruble, Applications Engineer
Introduction
Although HV91XX family used control single-switch converters topology size, their primary usage low-cost, medium power, discontinuous-mode flyback converters. Designing such converters relatively simple quick basic understanding flyback converter functions. purpose this note provide such understanding, illustrate, with couple examples, which such converter design proceed. should noted that this engineering approach, meant allow user develop working design quickly, textbook approach meant teach underlying theory. Safety margins taken into account, path taken through design intended make these margins work with each other order generate economical producible power supply. Many apparently arbitrary values used. They arbitrary, different ones could have been used that would have resulted different power supplies, that would have been, whatever feature optimized, just valid examples chosen. "The voltage main switch twice input voltage" incorrect, because voltage reflected from output winding either higher lower than input voltage (generally lower) depending voltage output, time allotted output inductance discharge into Discontinuous-mode operation merely means that energy (neglecting losses) into coupled inductor during time period when main switch then emptied during following period when main switch OFF. energy carried forward subsequent cycle. (See Figure both converter magnetic design, flyback magnetic thought independent inductors which share common core. Once designer accustomed thinking flyback magnetic dual inductor, rest design becomes easier. What designer needs define output side inductor that delivers enough energy load, while switch off, produce desired current desired voltage. Next, define input side inductor that takes enough energy when switch provide both output system losses. facilitate this, conversion formula necessary: 3tperiod This formula converts peaks noncontiguous triangle waves.1
Flyback Function
flyback converter functions, does almost every other switchmode converter, storing energy inductor during main switch period, then discharging stored energy into load during switch's period. trickiness this there any) that inductor more windings, input winding more output windings) that current flow alternates between input output windings, with effectively current (other than little leakage) flowing nonconducting winding while partner carries current. flyback converter works lead some confusion designer tries approach design magnetic were transformer, because, except case multiple output windings, magnetic flyback converter transformer. Perhaps easiest view magnetic flyback converter energy bucket which alternately filled (when main switch dumped (when switch OFF). flyback magnetic transformer despite superficial resemblance one: transformer functions voltage-in, voltage-out power transfer device, where input output windings conduct simultaneously. flyback magnetic energy-in, energy-out power transfer device where input output windings conduct current simultaneously. Obviously, voltages present active winding reflected, turns ratio, inactive winding,
deal with time percent total clock period, (duty cycle) define tperiod tperiod formula reduces input side output side Because designer knows length time switch will (these defined clock frequency used) well input output voltages desired, peak currents found from formulae used with defining formula inductance (dI/dt) determine required inductances input output sides coupled inductor. process, rest design generally falls into place.
HV91 Applications
Data Needed Start Design
Minimum Maximum Input Voltage Nominal Output Voltage(s) Tolerance(s) Maximum Output Wattage Minimum Output Wattage Maximum Allowable Output Ripple defined clock frequency list mechanical thermal constraints any)
reduced size vendor-dependent minimum, further reductions size will raise cost magnetics. Very small cores ultra-fine wire hard handle. There another important consideration choice frequency that often overlooked: dynamic range. difference between widest pulse generate (which function operating frequency) narrowest pulse produce function IC's speed internal structure) small, ratio between Pout(max) line Pout(min) high line must also small, reducing size inductor output filters will paid increasing size cost filter. Further, selected cannot handle full range pulse widths required, will start cycle-skipping (failing turn some cycles). While most ICs, including HV91XX family, simply skip cycles turning all, differential between Pout becomes great, skipping cycles reduces effective clock frequency converter, redefines minimum frequency which input filter must designed. example, converter skips every other cycle high
Operating Frequency
Most designers approach converter design with idea operating highest possible frequency that convenient. This generally useful approach, because minimizes size cost output capacitors coupled inductor. However, always best choose frequency. case low-power converters, once magnetics
"Energy Bucket"
IOUT
Situation varies, Load fixed. Slope rise dependent input inductance (fixed) (varying).
holds (LIIN) constant shutting input switch required current regardless long current took rise that level.
TON: Input side switch inductor charges
Slope fall dependent output inductance (fixed) VOUT.
IOUT TOFF: Energy stored previous half-cycle discharges into load varies (LI2) varying shutoff current allow more energy larger loads.
Situation fixed, Load varies.
IOUT Load
Output slope constant long output voltage constant. Dead time exists Output amplitude changes with output current. each cycle except when Load
Clock starts each cycle.
3-44
Figure
Load
HV91 Applications line/light load, size (and cost) filter doubled. Cycle skipping also increases either size output capacitors amount ripple converter's output. Recently, dynamic range been overlooked because most bipolar PWMs have wide dynamic range. example, bipolar 1845 operating 50KHz dynamic range only 17.6:1. CMOS 9110, contrast, dynamic range >120:1 50KHz. Proper dynamic range have significant effect filter cost. Another consideration choosing optimal frequency switching power loss, which increases linearly with frequency nonresonant converters. HV9110, which will accept input voltages 120V used. previously noted, this chip will allow dynamic range sufficient handle stated line/load variations 50KHz. Setting clock frequency requires selecting appropriate timing resistor. From graph data sheet, appropriate resistor 50KHz operation 330K. This however does account tolerance either resistor chip. ensure that device-resistor combinations operate above 50KHz, 261K better choice. reason that clock frequency should minimum rather than nominal value, despite reduction dynamic range this causes, prevent slowest converter from saturating coupled inductor. While magnetic saturation does cause damage current-mode converter would voltage-mode converter, still causes additional dissipation stress main switch. also limit power throughput.
Example
converter patterned after instrument power supply. This will simple generic example with bells whistles. First, need input parameters listed above: Maximum Input Voltage: Minimum Input Voltage: 65VDC 18VDC
Design
First, translate current major output winding maximum load peak current. From data sheet HV9110 seen that maximum time cycle minus approximately nsec. 50KHz, this amounts little over 49%. declare maximum duty cycle allow small safety margin. DMAX .49, then minimum .51. Using value (thus allowing overall dead band safety margin) determine peak secondary current: Ipk(5V) 8.0A
Outputs: 5.0V, ±1%, 0.25 25mV ripple 12.0V, ±5%, 0.01 0.7A, 0.5V ripple Maximum Output Wattage: 48.4W Minimum Output Wattage: 1.37W Operating Frequency: 50KHz
(See Figure
19.59A
+VIN
+12V
FDBK
VREF BIAS
3-45
COMP
HV9110
Figure
HV91 Applications This value will also used determine actual voltage required winding, which output voltage plus drop output diode: Vwinding Voutput F(diode) Also, repeat this procedure each auxiliary output winding determining real voltage required these windings: Ipk(12V) 0.7A voltage across resistor should slightly below 1.0V. establish maximum peak voltage that much less than will increase distance between maximum normal operating current maximum guaranteed overcurrent trip point, which 1.4V. Usually best choice operate with normal peak voltage across current sense resistor very close 0.99V reasonable value. Note that this voltage drop across current sensing resistor only occurs during current limit. normal operation loads below maximum, trip point switch moves down limit energy going output. That this form converter regulates. drop across switch more complicated, because first have choose main switch. this need estimate what current will Generally close enough estimate made using wattage into magnetic estimated minimum voltage across winding. assume that voltage across switch will greater than 1.5V peak, subtract this voltage, current sense voltage, from minimum input voltage estimate input current: (1.5V 15.5V Dividing previously determined input wattage inductor produces input current: 55.93W
1.715A
Next calculate minimum toff, which will maximum oscillator frequency. Using 261K timing resistor, maximum frequency should 67KHz, which gives tperiod 14.9 µsec, toff(min) 7.46 µsec. Next, need generate estimate instantaneous forward drop diode main output. cannot actually choose diode until know what reverse voltage needs which will known until input side inductance calculated. 0.8V should reasonable estimate. This voltage added output voltage determine actual voltage main output winding (5.8V this instance). Knowing peak current voltage output winding minimum toff, calculate inductance output winding from defining equation inductance, dI/dt. 5.8V
15.5V 3.61A
19.6A 7.46
2.21µH
Dividing that peak conversion factor (based 49%) gives estimated peak current. 3.61
same procedure used calculate primary inductance. First need calculate total power into magnetic. This power magnetic, plus losses magnetic itself. power magnetic just continuous output power converter, plus losses output diodes. average forward drop diodes this step. Pout(5V) 5.6V 8.0A 44.8W Pout(12V) 12.7V 0.7A 8.89W Pout(TOTAL) 44.8W 8.89W 53.69W losses well-designed magnetic assembly, fixed frequency power output, interestingly enough, depend primarily physical size magnetic. Smaller magnetics will less efficient hotter. Larger magnetics will have less loss cooler. effect logarithmic, means that will ever build magnetic less than efficient, because insulation required would burn under normal operating conditions, that very people could accept magnetic that over efficient, because size would prohibitively large. switchmode converter type described efficiency will result reasonable magnetic size that economical build. Using estimated magnetic efficiency 96%, calculate input power (which just output power divided magnetic efficiency): 53.69W
8.93A
Dividing previously assumed drop across switch estimated peak current gives target RDS(on) main switch: 1.5V
8.93A 0.168
Estimating that 100V should sufficient maximum drain voltage gives wide variety devices from which choose. IRF530 0.16 MTP20N10 0.15 closest.
Aside
Obviously, altering estimated value drop across switch down could change which switch ended satisfying circuit requirements. result does change design process, only efficiency final converter, much pays switch. also possible start design with main switch already selected mandatory efficiency goal, just fill appropriate value RDS(on) when that step process, doing mean repeating that section calculations once twice. Similarly, specify different transformer efficiencies, efficiency volume more than ordinarily important. Readers cautioned, however, that magnetic efficiencies below over result practical design. using IRF530, original estimate close enough there need recalculate, already calculated peak current determine value current sensing resistor needed, which simply Ipeak
55.93W
Next, determine minimum voltage across input winding. This just minimum input voltage converter, minus drop across switch current sensing resistor. drop across current sensing resistor easy determine from data sheet 9110. According data sheet, minimum trip point current limiting section chip 1.0V. This means that maximum normal operating peak 3-46
current sense, 0.99V
8.93A 0.111
closest lower value 0.110 0.11 values could also used with small risk worst-case combination causing current limiting less than 100% normal output.
HV91 Applications determine wattage input current: 3.61A2 0.11 1.43W 1.5W resistor would work. choices Motorola #MBR1045 #MBR1645, difference between them being that larger would more efficient. Similarly, when main switch OFF, addition present from input, there will 5.8V 2.40 13.9V reflected from output, total 78.9V present drain main switch. This leaves margin spikes. 100V should work. similar procedure based turns ratio finds voltage present diode output. This output ratiometrically linked winding with turns ratio 12.8 5.8, 2.20:1, when there 27.1V reflected from input winding there will 27.1V 59.6V, plus from output, total 71.6V across diode. 100V, ultra high speed silicon diode, like Motorola #MUR105 reasonable choice. Note that case multiple outputs which conduct same time, flyback magnetic does like transformer, this only case which does. Next, complete definition magnetic assembly. inductances input output side known. What remains define resistances windings. These calculated from rule that optimum size/efficiency magnetic, loss occurs resistive loss windings, this loss balanced among windings based percentage total power handled each (the other loss occurs core hysteresis loss). Output power, previously calculated, 53.69W. Input power calculated 55.93W. Thus power loss magnetic 55.93W 53.69W 2.24W. copper loss should close 1.12W. Half this, .56W, occurs input winding, which must supply outputs. other half split between windings ratio their respective powers. winding this
Word Current Sense Resistors
Obtaining good current sensing resistor still problem. Most common resistors this service because they inductive. What answer there probably lies bulk metal resistors, noninductive resistors, careful. Some "noninductive" resistors only "noninductive" frequencies, source considerable error high frequencies. Carbon film resistors most metal film resistors recommended. Also, most value resistors that look like carbon composition resistors actually film wirewound resistors molded cases. 4-terminal resistors specifically meant current sensing most part wirewound, meant only switched current measurements. sure test inductance resistor intend before install your circuit! Also, even good noninductive resistor will work properly long leads long printed circuit board traces allowed inductance mechanical assembly. Good layout practice mandatory.
Design
(continued)
also same procedure determine approximate power loss main switch. This absolute loss, which will little higher rise RDS(on) with temperature MOSFET, generally will close enough start determination heatsinking requirements. 3.61A2 0.16 2.085W Note that because input current equivalent this instance current through switch current sense resistor), does need account duty cycle time effects. Next, need determine inductance required input winding. that know voltage across winding peak current through winding, need calculate minimum repeat same procedure output: 14.92µsec 7.31µsec then 15.5V
44.8 53.7
.467W
winding 8.89 53.7 .093W
Knowing target wattage current each winding this case RMS) calculate resistances from input: .56W
8.93A 7.31µsec
12.7µH
(3.61A)2
output: .467W output:
that have inductances output input windings determine voltage stresses applies switch diodes, make final determination appropriate devices. trick here that inductance varies square number turns, turns ratio varies square root ratio inductances. turns ratio
(8A)2 .0073 .093W (0.7A)2 0.190
This completes definition magnetic assembly. Actually, because difficult balance power loss between windings, between windings core, easing calculated values somewhat much 20%) result magnetic that would significantly smaller with increase total losses. This should discussed with your magnetics vendor. Also, because modern high-performance ferrites tend have very losses moderate frequencies like 100KHz, wish divide total power loss differently, core, copper. This also reduce cost inductor without increasing size. This probably will work clock frequency converter 200KHz more. 3-47
12.7 2.21
2.40 .417
Thus, when there present input winding, there will .417 27.1V output winding. Adding this that will present cathode diode from output gives 32.1V, means that diode allows margin noise spikes should work well. good
HV91 Applications
Leakage Inductance
final thing need specify with regard magnetic maximum leakage inductance. Leakage inductance measure amount flux generated winding magnetic assembly that coupled other winding(s) core winding structure. flyback converter measure much energy taken into input winding incapable being transferred output winding when switch turns OFF. This energy appears voltage spike drain MOSFET each time turns must dissipated either MOSFET directly, snubber circuit. reasonable value leakage inductance nominal inductance, this highly variable depends intended operating frequency, size, efficiency magnetic being developed. actual maximum value should discussed with your magnetics vendor before cast concrete, that maximum value should used later development snubber, snubber appears worthwhile. (See Figure Next, select output capacitors. criteria need met. First, minimum capacitance must satisfy standard capacitance definition dV/dt where Amperes, Farads, delta delta allowable output ripple. Second, almost inevitably harder, Equivalent Series Resistance (ESR) capacitor(s) must provide more than part ripple (75% this case) provided from first criteria, accordance with Eripple Ipeak where Ipeak peak current from output inductance during discharge. (This because when main switch turns OFF,
current filter capacitor switches, effectively instantaneously, from outbound current Iout, inbound current, (Ipeak Iout). reason splitting allowable ripple between criteria that final converter they will tend add. reason asymmetrical split allowable ripple that ESR-caused ripple limit more difficult criteria meet. some instances more drastic partitioning more favor ripple) better. output these criteria calculate follows: (.75 .025V)
((.25 .025V)/10µsec) 12,800µF
19.59A 957µ
Based Mallory type capacitor (330µF, WVDC, 0.04) typical good output filter capacitor). This works pieces parallel satisfy ripple from standpoint, pieces satisfy ripple from capacitance standpoint. Close enough. Note that capacitor chosen tantalum capacitor, wish aluminum capacitors perform same service ignore ripple from capacitive droop assign 100% ripple ESR. Sizing aluminum capacitor strictly from will generally provide with times more capacitance than needed. This will slow down transient response converter, means that will rarely, ever, encounter stability problems.
Aside
Based price Mallory THFs, stated solution optimum solution problem. better solution might change effective ripple specification from 0.025V 0.250V additional stage filtering from nominal output "real" output seen load. This means that instead capacitors these must followed filter with 10:1 attenuation (20dB) 50KHz. This implies corner frequency, 5KHz, which means won't small filter, there necessity using high performance capacitor this second filter stage. other difficulty with second-stage filter that resistance inductor cancelled feedback loop, consequently variation output voltage with load current exceed specifications power supply. hold regulation specification line would need inductor with .003 resistance. second alternative would combination electrolytic film capacitors parallel with electrolytics sized solely load current ripple criterion film capacitors sized solely ripple criterion. this instance, capacitive droop should divide ripple about 50-50. (See Figure Next define filter capacitors output same way: 0.7A
Coupled Inductor Specification
(Preliminary)
(input)
(outputs)
Schematic
Nominal Operating Frequency: 1-2: 12.7µH with Flowing 0.045 Leakage Inductance with shorted: 200nH 3-4: 2.2µH with 19.6 Flowing 0.0075 3-5: Voltage ratio 12.8: 0.19 Polarization: Starts must shown schematic (pins Insulation: Vacuum impregnate class thermosetting varnish Interwinding Insulation: applicable Expected Thermal Rise: <45°C 40°C ambient Mounting: Through-hole
therefore 56µF 10µsec
(.75
1.715A, therefore 0.219
This much easier capacitor find. Sprague type 676, 900µF 12WVDC, aluminum electrolytic (the smallest capacitor this series) will work well. 56µF, 15V, Mallory type will also work.
Figure 3-48
Next define divider resistors which will used feed back sample output error amplifier PWM.
HV91 Applications
Output Filters Equivalent Ripple
pcs. Mallory #THF337M006P1G Total 13,860µF Cost: Highest Response: Fastest Volume: Smallest
Mylar
Feedback
pcs. Mallory #THF337M006P1G United Chemi-Con #RZA 22,000µF 6.3v 1.5µH; .0025 Total 23,320µF Cost: Intermediate Response: Intermediate (Reduced Load Regulation) Volume: Largest
1.55µH, .0025 22,000µF Mylar
Feedback should taken from before second filter avoid stability problems.
pcs. Sprague #676D159M6R3JT5C Total 60,000µF Cost: Least Response: Slowest Volume: Intermediate
Mylar
Because error HV9110 CMOS, input bias current negligible, divider string carry very small current; 100µA plenty. feedback terminal HV9110 (the inverting terminal error amplifier) satisfied 4.00V ±1%. 100µA divider, lower resistor should 4.00V
Feedback
Figure
Modeling, Analysis Design Converters, vol. VPEC staff, VPEC2, ISBN (none)3 Advances Switched-Mode Power Conversion, vol. Middlebrook Teslaco, ISBN (none)4 Dynamic Analysis Switching-Mode Converters Nathan Sokal, Andre Kislovski, Richard Redl, Nostrand Reinhold, ISBN 0-442-21396-4 Modern Switchmode Power Converter Circuits Severns G.E. Bloom, Nostrand Reinhold, ISBN 0-442-21396-4
.0001A 40200
closest real value 40.2K. produce exactly 4.00V with 40.2K need actual divider string current 4.00V 99.50µA
Dividing remaining between intended output actual divider string current, gives value 1.00
.00009950 10,050
Accessory Circuits
snubber circuit MOSFET drain switching spike should sized absorb energy taken into magnetic that coupled output. available energy where leakage inductance primary peak input side current. Using reasonable estimate leakage inductance gives value 250nH. Spike energy then (250nH 8.93A2) 10.1µJ Multiplying this maximum repetitions second (which occurs maximum frequency) gives 10.1µJ 67,000Hz .679W, which amount power dissipated either MOSFET snubber. dissipate snubber must captured snubber capacitor without exceeding drain breakdown FET. Minimum breakdown: 100V Maximum circuit-supplied voltage drain: Maximum voltage snubber cap: 3-49 21.1V 78.9V
closest value 10.0K. Normally, next final step design process would stability analysis. However, turns that advantages discontinuous-mode, current-mode flyback converters with maximum duty cycle 50%, with output capacitors (especially they aluminum electrolytics) sized ripple requirements, that they usually stable is." matter prudence, checking loop response power supply network analyzer Venable machine always good idea, supplies this nature this writer longer considers full analysis mandatory. this reason, because including stability analysis this application note would probably double length, analysis omitted. those desirous performing full mathematical analysis every loop, following texts5 subject recommended:
Switching Regulator Analysis Mitchell, McGraw-Hill, ISBN 0-07-042597-3 Switch Mode Power Conversion Sum, Marcel Dekker, ISBN 0-8247-7234-2
HV91 Applications
High Switched Current Paths
+VIN
+12V
Output Loop: 19.5A delta Input Loop: 8.9A delta Output Loop: 1.7A delta MOSFET drive Loop: 0.6A delta MOSFET drive Loop: 0.6A delta
FDBK
VREF BIAS
COMP
calculate size snubber capacitor convert energy previously calculated energy, divide maximum voltage desire capacitor: 10.1µJ)
Figure
21.1V2 0.045µF
Using next larger real capacitor (.047 this case) assures that voltage spike will large enough break down MOSFET. resistor series with snubber capacitor must have enough value allow capacitor discharge minimum on-time switch, which HV9110 will about 200nsec. Because discharge distance (62.5V max) much greater than charge distance (21.1V) declaring 400nsec will work. Thus 400nsec
prematurely. time constant this filter should approximately snubber time constant, never more than nsec. (Otherwise authentic fault current spikes slowed down much.) current spike filter larger, much smaller, because load presented HV9110 quite small order 3pF). Using resistor 75pF capacitor should sufficient snubber above. small capacitors shown Fig. connected directly converter outputs. These stacked film capacitors with very good characteristics, intended general noise suppression. They necessary, they reasonable insurance. Input filter will also required under most circumstances. conducted emissions, generally asymmetrical pi-type filter sufficient. converter-side capacitor should sized convert delta caused switching reasonably delta voltage over toff. inductor input side capacitor should designed have corner frequency that complements corner frequency regulator loop, minimize susceptibility outside noise coming regulator. this case, from previous calculations, input switching current known maximum 8.9A. Similarly, minimum toff 7.46µsec, reasonably value delta 250mV. Thus, from dV/dt, converter side capacitor calculates 8.9A 3-50
47nF 8.51
catch here that, except when switch minimum time, capacitor will reverse charge (VFET current sense) 62.5V. this energy which must dissipated resistor: (62.5V2 47nF) 67KHz 6.15W resistor will necessary. save 679mW FET, this hardly seems worthwhile, done desired. snubber used, filter network should added between current sense resistor current sense terminal HV9110 prevent higher-than-usual leading edge spike current waveform from shutting switch
(.25V 10-6 sec) 271µF
HV91 Applications good choice 330µF. This capacitor also serve insure minimum holdup time short input dropouts. values inductor input side capacitor calculated from current example, dI/dt secondary will over amps microsecond! This sufficient cause significant radiated improperly handled. Controlling from switchmode converter neither difficult costly, provided attention paid subject early enough design cycle. There "tricks" control, only basic rule: Minimize area loops around which switched currents circulate. This purely mechanical constraint, should dealt with during layout. Generally, unless have layout person with prior experience with switching converters, circuit designer will have lead board designer through first layouts, even thereafter, will have show board designer where switched current loops circuit. (See Figure Obviously, there other constraints switchmode converter well, these also affect system performance. Most converters should laid single-sided, with second side reserved ground plane. Also, high currents require wide lands, just keep resistances low. First foremost though, should effort keep switched current loop area minimized. Loop length also important, long runs have enough inductance disrupt circuit operation. 9110/9120 this only likely around gate drive loops, which have relatively delta I's, current sense resistor, where stray inductance cause overcurrent sensing shut main switch prematurely, thereby limiting power output.
once corner frequency regulator loop known. this case regulator loop rather slow, large output capacitors required ripple specifications, appropriate frequency only 750Hz. This gives 10-6 This means that combination inductor capacitor values whose product 10-8 will provide adequate filter. Generally, best larger value capacitor smaller value inductor because inductors usually cost more. this case using 1000µF capacitor results 45µH inductor, 50µH easily obtainable, small inexpensive. should enough that previous calculations based remain valid. Based 3.6A maximum input current, will give 0.14V drop, which well within allowable safety margins, results cheap, small inductor.
Layout Noise (Radiated EMI)
Anyone using HV9100, HV9110, other parts same family, will switching current flow off. Sometimes, quite large currents switched short periods time.
Effect Local -VSS Capacitance Area MOSFET On-Drive Loop
+VIN +12V
Current loop with local capacitor Area deleted with local capacitor Resultant smaller loop
FDBK
VREF BIAS
COMP
3-51
Figure
HV91 Applications Board layout should proceed taking switched current loops order delta laying portions circuit last. Using first circuit example, this means starting with loop, (which includes only coupled inductor, diode, output capacitors) taking input loop next (coupled inductor, power MOSFET, current sense resistor, input filter cap) following those with output loop, MOSFET drive loops. each instance, entire loop that matters, including return path. Assuming that return path good, even boards with ground planes, risky. Look each loop carefully that area minimized. After switched current loops laid out, sections fitted where they convenient. They contribute noise, they convey generated elsewhere. feedback loop special case. itself, this loop, susceptible noise generated other loops, because most part, high-impedance path, much energy required disrupt feedback loop should also laid minimum area, more important that path circuitry lies well away from, where possible perpendicular switched current loops. Generally, layout grows outward from transformer, careful choosing which pins transformer connect which windings make layout convenient. There things that done designs with HV91XX family that make specific layout more convenient: First, should remembered that high current paths associated with current sense resistor include line from junction resistor MOSFET current sense terminal sense lead very current path, comparatively long providing that path from MOSFET resistor ground short. output lead from HV9110s HV9120s however should kept short, because services both charge discharge paths from MOSFET gate. Also, when winding transformer used power HV91XX, help split filter capacitor into pieces, near transformer diode keep that current loop short) second near HV91XX keep ON-drive current loop short. terminals HV91XX adjacent each other specifically allow this. (See Figure Inevitably, there will some residual radiated EMI, some this will picked circuits seen input outputs conducted EMI. film capacitor previously noted should suffice remove this from outputs, small commercial line filter should suffice input. Most commercial filter suppliers offer services (sometimes free!) assure that converter filter meets whatever requirements force your particular circumstances. Using these services final check generally worthwhile.
Final Design
47µH, +VIN 1,000µF 100V 330µF 100V IN4148 +12V MUR105 56µF
MBR1045 15,000µF 6.3V
10.0K 261K
IRF530
FDBK
0.1µF 40.2K 75pF 390K
VREF BIAS
COMP
HV9110
0.111 Noninductive
Figure
3-52
HV91 Applications
Example
+VIN 390VDC Option ISO1B
Output return
FDBK
VREF BIAS
COMP
HV9120
converter's outputs exit enclosure generally used, replaced with bypass caps loads they supply.
ISO1A
Option
Input return Return
Figure
There only additional noise control measure necessary. reference HV91XX high impedance node, designed work with 0.01µF 0.1µF capacitor between itself VSS. This capacitor should omitted. other hand, capacitor between bias should required. capacitor from bias improves operation, capacitor should placed from VSS. usual switching circuit, noise filtration capacitors should types with good high frequency performance: Stacked Mylar ceramic multi-layer caps generally best. (See Figure
Design
First, select timing resistor. assure that units operate 500KHz above despite tolerance effects, 16.5K resistor should sufficient. But, because parasitic capacitances associated with layout have serious effect clock oscillator performance high frequencies (the timing capacitance 9120 totals less than 10pF), value timing resistor should confirmed final assembly assure desired performance. Next, calculate minimum toff ton. minimum frequency 500KHz, worst case maximum 600KHz, 600KHz, maximum duty cycle will greater than 46.5%. Thus, minimum will 1667 nsec .465 nsec minimum toff will 1667 nsec 53.5% total period. peak current calculations follow) will sufficient declare (duty cycle) 46.5%, 51.5%, leaving deadband assure discontinuousmode operation. Next, translate current output with greatest percentage load (+5V this time) peak current using same formula previous example: .55A
Example
converter DPM. Size small consistent with cost, input range 65VAC 240VAC, load fairly constant. before, first thing need specifications design Maximum Input Voltage: 390VDC {[(240 15%) 2]-1.4} Minimum Input Voltage: 90VDC [(65 2)-1.4] Outputs: 0.55A, 100mV ripple 30mA, ripple +10V 10%, 14mA, 100mV ripple Maximum Output Wattage: 3.04W Minimum Output Wattage: 2.29W Operating Frequency: 500KHz (min)
.515
1.328A peak
HV9120, which accepts input voltages 450VDC will required. This time, even with high operating frequency, sufficient dynamic range exists (13.2:1) that supply should exhibit cycle skipping. This minimizes size output filters. (See Figure
This allows calculate inductance secondary using dI/dt. Remember that voltage seen inductor includes forward drop output diode, actual calculation works 5.75V
1.328A 858nsec) 3.72µH
HV91 Applications winding will equal turns (thus also voltage inductance) winding. winding, (which powers HV9120) conducts same time main winding, thus turns ratio equal voltage ratio, (10.7 5.7) inductance calculation necessary. SIDE NOTE: load stated above winding (14mA) considerably larger than specification HV9120. remaining 13mA what required provide charge gate power MOSFET HV9120 will driving 500KHz. This current determined dividing total gate charge MOSFET (Qg) (from MOSFET data sheet) (Vg) determine effective gate capacitance, then using that capacitance value CV2f (converting charge current) determine current required charge gate 500,000 times second. While this calculation good check real supply current HV9120 operation, seldom necessary unless converter operating over 100KHz. Next, using same system, calculate required inductance input winding. start, need power into magnetic, which just power magnetic divided efficiency. Power magnetic includes only output power, voltage drop through output diodes. currents needed, 0.75V safe estimate diode drop. maximum power magnetic will [5.75V (.55A+.03A)] (10.75V .014A) 3.485W Because more difficult design small magnetics (and size original constraints) high efficiency, because higher frequency magnetics tend less efficient, this time will adopt efficiency estimate only 94%. Now, power into magnetic calculates 3.485W
Coupled Inductor Specification
(Preliminary)
Schematic
Nominal Operating Frequency: 1-2: 625µH with 0.10 Flowing Ohms Leakage Inductance with shorted 10µH 3-4: 3.7µH with Flowing 0.16 Ohms 5-4: Voltage ratio 1.00 1.00 Ohms 6-7: Voltage ratio 10.7 12.5 Ohms Polarization: Starts must shown schematic (pins Insulation: Vacuum impregnate with class thermosetting varnish Interwinding Insulation: 3-4-5 1.5KVAC 500VAC Expected Thermal Rise: <60°C 50°C ambient Mounting: Through-hole
0.94 3.707W
obtain input current, this wattage divided minimum voltage across input winding, which just minimum input voltage less drop current sensing resistor power MOSFET. maximum drop across current sensing resistor again should just under 1.0V (from HV9120 spec.) reasonable estimate drop MOSFET 2.1V, (based Supertex #VN0660N3, 600V, MOSFET). minimum input side voltage will 2.1) 86.9V input current will 3.707W
Figure
86.9V 42.7mA
Knowing input current duty cycle calculate peak input current, which will 0.0427A
copper loss 111mW. This should divided among various windings proportion their proportion total wattage. Input winding: 50%, 55.5 output: 45.4% 50.4
.465
0.108A
output: 2.5% 2.75 +10V output: 2.2% 2.40
Knowing peak input current, minimum input voltage smallest maximum ton, calculate input side inductance from dI/dt. This works 86.9V
.108A 775nsec
624µH
Actually, this shortchanges input winding somewhat, carries slightly more power than outputs combined, this usually trivial. before, current (NOT peak) used determine resistance knowing dissipation: input winding: .0555W winding: winding: winding: 3-54
Next, need determine resistances various windings magnetic. this case, because operating frequency, copper losses core losses probably will approximately equal. Again, power loss magnetic just Pout 222mW. Assuming half this copper loss gives
.0427A2 .0504W .55A2 .166 .00275W .030A2 3.06 .0024W .014A2 12.24
HV91 Applications This gives enough information complete specification coupled magnetic. (See Figure value current sensing resistor determined once peak input current known. sensing voltage levels HV9120 same they were HV9110 example Thus 0.99V added present output capacitors. Schottky diodes, such 1N5819, should work well output diodes. increased efficiency were required, 1N5822 three-amp Schottky diodes, and/or 500V, VN0650N3 main switch could substituted. with first example, next thing calculate requirements output filter capacitors. This time ripple specification easier meet (100mV loads smaller. technique remains same, 25%/75% division ripple between capacitance should still hold. Thus, output 0.55A
0.108A 9.16
next lower resistor 9.09 last time, using resistor will probably result very any) units that allow full output power line. Wattage current sense resistor calculated using input current (42.7 mA). This calculates 16.6 based resistor, 1/10 watt resistor find noninductive one) used. drop across main switch power loss should calculated next. this case already selected main switch Supertex VN0660N3) based solely being smallest (and least expensive) 600V MOSFET available. on-resistance VN0660N3 which implies peak-current voltage drop 2.16V, power dissipation 36mW, which easily handled TO-92 version. Using "square root inductances ratio" determine approximate voltages reflected across coupled inductor determine actual voltages present main switch when diodes when they blocking. This time that works almost exactly 13:1. Thus when main switch off, will see: 5.7V added 390V present from input, total 464V. diodes, when blocking, maximum
(.025V 10-9sec) 20.5µF
Remember that capacitor holdup, maximum rather than minimum used, time which capacitor must hold maximum time switch could possibly ripple also done exactly before: .075V
1.328A .565
This much easier capacitor find. Nichicon type 150µF, 6.3V would work fine. would Sprague 672D227H6R3CG3C (220µF, 6.3V aluminum) Sprague 199D336X96R3DA1 (33µF, 6.3V dipped tantalum). secondary works same way, except current only 30mA: 0.030A
10-6
(.025V 10-9sec) 1.1µF .075V .073A
this case, because load current (and thus capacitor) small, probably better stacked film capacitor ignore which will orders magnitude below requirements. Wima #MKS-2 (ESR .02) would fine. same capacitor could also used output that feeds HV9120 (14mA). This particular capacitor also excellent choice final noise filters
Final Design
+VIN 390VDC Option 71.5K 150K Note 16.5K
33µF 6.3V
1N5819
Output return
1N4148
1N5819
VREF BIAS
4.7M 100K
4700pF
Input return
220pF Note 0.1µF 33Pf 100K Note 390K
150K
150K
FDBK
COMP
HV9120
VN0660N3
converter's outputs exit enclosure generally used, replaced with bypass caps loads they supply.
6.19K ISO1A
Option
9.09 Noninductive
4N26
Note Delete with Option
6.19K Return
TL431CLP
Figure
3-55
HV91 Applications regulator's outputs example this case regulator's outputs will outside enclosure extra noise filters probably unnecessary. Feedback this circuit shown ways: First, just resistive divider secondary that feeds HV9120, second optical feedback, which requires additional circuitry. (See Figure 10.) regulation specifications given, (±5%) resistive divider separate winding sufficient over industrial temperature range (-40°C +85°C). Such "divider separate output" relies magnetic coupling between windings common core regulate isolated windings. Within limits (accuracy, mostly) works very well, would require excellent transformer builder able meet regulation magnetically coupled outputs over full -55°C +125°C military temperature range. resistive divider, previous example, draw very little current, because CMOS error HV9120 does draw significant bias current. Because last example used 100µA divider current this will 40µA. (Using much less than 20µA requires using high value precision resistors that expensive.) Again, feedback point HV9120 internally trimmed expect 4.00V design output voltage. This time design voltage winding directly coupled divider 10V, using 40µA divider current, lower resistor becomes 100K, upper resistor becomes 150K. original feedback divider HV9120 replaced divider composed phototransistor emitter load. Precision resistors this divider longer required because regulation loop will compensate errors here. current used earlier this path (40µA) kept, adjusted convenient. will stay with original value. choice optocoupler will depend loop response speed required. Optocouplers tend slow, unless care taken optocoupler selection, optocoupler ends being controlling element regulation loop response speed. this example, used 4N26 because hand. 6N135 similar high speed optocoupler would have given loop response more appropriate 500KHz converter. resistor shown between base optoisolator transistor ground noise/leakage eliminator should have value between 10M, depending optocoupler used. 4N26, 4.7M works well. Because there amps regulation loop, loop gain will more than necessary, some must done away with. Otherwise stabilization will problem. There simple ways eliminate gain. Either convert error HV9120 gain configuration with equal resistors, resistor between anode TL431 output return reduce gain approximately Both methods work equally well. error HV9120 minimum guaranteed output sink current 120µA, feedback resistor greater than will allow full opposite swing. 150K gives plenty margin, reduces power little. Alternatively, resistor between anode TL431 output return used gain-destroyer reduce gain TL431 (plus optoisolator) approximately value gain-destroyer resistor dependent coupling "gain" optoisolator current required from phototransistor. Using 4N26, current required from phototransistor approximately 40µA, current required achieve will approximately 100µA. adequately reduce gain TL431 will require delta anode about 50mV, resistor should work.
Accessory Circuits
snubber circuit shown this converter, none should necessary. Maximum energy available snubbed, like last time, just Lleakage Ipeak2, 117nJ switch-off. 600KHz that works 70mW, which easy ignore. Also, main switch chosen 20pF reverse transfer capacitance, which absorb this much energy while only rising additional 76V. This still leaves Vdrain MOSFET below breakdown should safe. Leading-edge spike suppression current sense resistor handled first example, with resistor between current sense resistor current sense HV9120, plus capacitor between current sense terminal ground. this regulator, leading edge spike suppression probably more important than last one, because peak gate drive current power MOSFET actually greater than load current! Because gate capacitance much smaller though, capacitor's size should reduced. 33pF good starting value. Optical feedback usually only used isolated outputs that must regulated tighter tolerance than over industrial temperature range ±7.5% over military temperature range. Optically isolated feedback been developed over past years that straightforward relatively inexpensive, consisting T.I. #TL431, optocoupler four resistors. Because high gain TL431 (80dB), virtually level accuracy desired achievable. TL431 requires 2.5V third terminal achieve regulation. require output, divider resistors will equal. TL431 requires maximum into reference terminal. hold reference current maximum divider error, divider current must 400µA. Thus divider resistors should 6250 each. reasonable value 6.19K. 3-56
Conclusion
demonstrate functionality, both examples were assembled tested technician facility. suggestions avoid difficulties follows: First, wire-wrap construction methods are, always will completely incompatible with power supply construction. light gauge wire will carry current, stray inductance caused longer-than-necessary paths will disrupt circuit cause additional EMI. Seriously, requirement short, low-inductance, low-resistance paths good mechanical layout throughout design mandatory. Every unnecessary tenth inch lead should eliminated. This appear save space completed design, fact will save both space trouble. Second, flyback power supplies should never operated without load! Once main switch turns off, energy stored coupled inductor inevitably goes into output charges output filter capacitors. load present remove charge, capacitors output diodes will break down. Third, output voltage ratios multiple output windings need adjusted slightly, output voltages into
HV91 Applications tolerance. This happened small converter where output winding, because closer input winding than other output windings, more voltage than planned. solution reduce number turns that winding about 10%. Last, choice optoisolator 4N26) appropriate. result that regulation loop crossover frequency only 9KHz, when should have been over 100KHz. transistor with 10µsec storage time, like phototransistor 4N26, just isn't capable response speed desired from 500KHz switcher. summation, different circuits have been developed show flexibility flyback converters built with HV91XX family ICs, simplicity their design construction. Both circuits their original design goals. field HV91XX family broader than illustrated single application note. Many other forms converters, which best suited their particular purposes also built using HV91XX ICs. Contact Supertex additional application notes. Reference Data Radio Engineers, Howard Sams Co., chapter table Virginia Power Electronics Center, Bradley Department Electrical Engineering, Virginia Polytechnic Institute State University, Blacksburg 24061. Available from publisher footnote Only available from publisher: TESLAco, Mauchly, Irvine 92718 (714) 727-1960. Books with ISBN numbers ordered from bookstore. books this list except TESLAco book also acquired from: E.J. Bloom Assoc. Educational Division, Duran Dr., Rafael 94903-2317 (415) 492-1239. They generally have them stock. Magnetic assemblies converters were supplied Manufacturing, Inc. Crosby Rd., Dover 03820-1409 (603) 742-4375.
3-57
Other recent searches
SAM-4 - SAM-4 SAM-4 Datasheet
PS2845-4A - PS2845-4A PS2845-4A Datasheet
NLAS4157 - NLAS4157 NLAS4157 Datasheet
MJF15030 - MJF15030 MJF15030 Datasheet
MJF15031 - MJF15031 MJF15031 Datasheet
GP2Y0D340K - GP2Y0D340K GP2Y0D340K Datasheet
CFAG12864A-TMI-V - CFAG12864A-TMI-V CFAG12864A-TMI-V Datasheet
BSO150N03 - BSO150N03 BSO150N03 Datasheet
Privacy Policy | Disclaimer
© 2012 Datasheet Archive
