2-Phase, Single Output, Synchronous Buck Control MIC2155 MIC2156

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2-Phase, Single Output, Synchronous Buck Control MIC2155 MIC2156

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MIC2155/2156
2-Phase, Single Output, Synchronous Buck Control
MIC2155 MIC2156 family 2-phase, single output synchronous buck control featuring small size, high efficiency, high level flexibility. implement 500kHz (MIC2155) 300kHz (MIC2156) voltage mode control, with outputs switching 180° phase. result out-of-phase operation 1MHz 600kHz) input ripple with ripple current cancellation, minimizing required input filter capacitance. output voltage tolerance allows maximum level system performance. Internal drivers with adaptive gate drive allow highest efficiency with minimum external components. independent enable pins power good output provided, allowing high level control sequencing capability. MIC215x family junction operating range from -40°C +125°C. Data sheets support documentation found Micrel's site www.micrel.com.
Features
Synchronous Buck Control with outputs switching 180° out-of-phase Remote sensing with internal differential amplifier 4.5V 14.5V input voltage range Adjustable output voltages down 0.7V output voltage accuracy Starts into pre-biased output 500kHz operation (MIC2155) 300kHz operation (MIC2156) Adaptive gate drive allows efficiencies over Adjustable current limit with sense resistor Senses low-side MOSFET current Internal drivers allow phase Power Good output allow simple sequencing Dual enables with micro-power shutdown UVLO Programmable soft-start Output over-voltage protection Works with Ceramic output capacitors Multi-input supply capability Single output high current capability with master-slave current sharing External Synchronization Small footprint 32-pin MLF® Junction temperature range -40°C +125°C
Applications
Multi-output power supplies with sequencing DSP, FPGA, ASIC power supplies Modems Telecom Networking equipment Servers
MicroLead Frame registered trademarks Amkor Technologies Micrel Inc. 2180 Fortune Drive Jose, 95131 (408) 944-0800 (408) 474-1000 http://www.micrel.com
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156
Contents
Ordering Information Typical Application Configuration. Description Absolute Maximum Ratings Operating Ratings Electrical Characteristics. Typical Characteristics Functional Diagram Functional Description. Startup Soft Start Enable. Supply Voltages Internal References UVLO Power Good Oscillator Frequency Synchronization MOSFET Gate Drive Circuitry dv/dt Induced Turn Low-Side MOSFET. Remote Sense Setting Output Voltage. Current Limit Overcurrent Protection Current Limit Setting simple method Accurate method Example: Inductor Current Sensing. Current Sharing. Startup into Pre-Biased Output. Separate Input Supplies Component Selection, Guidelines Design Example Output Filter Selection Output Capacitor Selection. Inductor Current Sense Components Input Capacitor Selection. MOSFET Selection Current MOSFET Power Dissipation Calculation. External Schottky Diode Snubber Design Compensation Output Voltage Loop Error Amplifier Design Procedure Step Decide crossover frequency. Step Determine gain required crossover frequency Step Determine gain boost needed crossover frequency (fc) Step Determine frequencies Step Determine frequency fz1. Step Determine frequency fp2. Calculating Error Amplifier Component Values Compensation Current Sharing Loop General Layout Component Placement Design Layout Checklist Package Information
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156
Ordering Information
Part Number MIC2155YML MIC2156YML Frequency 500kHz 300kHz Voltage Adj. Adj. Junction Temp. Range(1) -40°C +125°C -40°C +125°C Package 32-Pin MLF® 32-Pin
Lead Finish Pb-Free Pb-Free
Typical Application
500µF
10µF FDMS7672 VOUT 1.8V 1.5µH
0.1µF
FDMS7672
0.22µF
FDMS7660
FDMS7660
250µF
2-Phase Converter
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156
Configuration
PGND1 LSD1 VIN1 LSD2 PGND2 BST1 HSD1 COMP1 BST2 HSD2 VIN2 VOUT COMP2 EA2+ PGOOD
Description
Number Name Function
BST1 HSD1 COMP1 DIFFOUT RMVOUT RMGND AGND AVDD SYNC
Boost1 (Input): Provides voltage high-side MOSFET driver gate drive voltage higher than source voltage minus diode drop. High-Side Drive (Output): High current output-driver external. high-side MOSFET. Switch Node (Output): Return HSD1 Current Sense (Input). Current-limit comparator noninverting input. Current sensed across side low-side during off-time. Current limit resistor series with pin. Enable (Input): Channel enable. Pull high enable. Pull disable. Soft Start (Input): Controls turn-on time output voltage. Active Power-up, Enable, Current Limit recovery. Compensation (Input): Output internal error amplifier Channel Feedback (Input): Negative input error amplifier Channel Output remote sense differential amplifier. Remote VOUT: Connect Vout remote sense point. Input precision differential amplifier. Remote Ground: Connect Ground remote sense point. Input precision differential amplifier. Analog Ground Analog supply voltage (input). Connect through filter network Connect Sync (Input) Synchronizes switching external source. Leave floating when used.
2009
DIFFOUT RMVOUT RMGND AGND AVDD SYNC
32-Pin MLF® (ML)
M9999-052709-A (408) 944-0800
Micrel, Inc.
Number Name Function
MIC2155/2156
EPAD
PGOOD EA2+ COMP2 VOUT VIN2 HSD2 BST2 PGND2 LSD2 VIN1 LSD1 PGND1
Power Good (Output): Asserts high when voltage rises above Power Good threshold. (input) Positive input Channel (current sharing) error amplifier. Connect Channel current sense. (input) Negatve input Channel (current sharing) error amplifier. Connect Channel current sense. Compensation (Input): external compensation Channel Output sense (input): Connect output side inductors. Used current sharing. (Input) Supply voltage Channel Used Channel UVLO circuit. Connect Switch node (Output): Return HSD2. High-Side Drive (Output): High current output-driver high-side MOSFET. Boost (Input): Provides voltage high-side MOSFET driver Channel gate drive voltage higher than source voltage minus diode drop. Power Ground High current return side driver Low-Side Drive (Output): High-current driver output Channel low-side external MOSFET. Internal Linear Regulator from VIN1 (Output): ext. MOSFET gate drive supply voltage internal supply When VIN1 <5V, this regulator operates drop-out mode. Connect external bypass capacitor. Enable (Input): Output enable. Turns both sides. Pull high enable. Pull disable. Supply voltage Channel (Input): Used UVLO circuits. Low-Side Drive (Output): High-current driver output Channel low-side external MOSFET. Power Ground High current return side driver Exposed (Power) Must make full connection plane maximize thermal performance package.
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156
Absolute Maximum Ratings(1)
Supply Voltage (VIN 1,2) -0.3V Bootstrapped Voltage (VBST) FB1, RMVOUT, RMGND, AVDD, Sync, EA2+, FB2, VOUT -0.3V CS1, EN1, -0.3V Junction Temperature Range.-40°C +125°C Ambient Storage Temperature.-65°C +150°C 1500V machine model 100V human body model Lead Temperature (soldering 10sec). 260°C
Operating Ratings(2)
Supply Voltage (VIN 1,2). +4.5V +14.5V Output Voltage Range. 0.7V 3.6V Package Thermal Resistance MLF® (JA) .50°C/W MLF® (JC).5°C/W
Electrical Characteristics(3)
25°C; VIN1 VIN2 =12V; unless otherwise specified. Bold values indicate -40°CTJ+125°C Parameter VREF Supply Total Supply Current, IVIN1 IVIN2 Mode Supply Current Shutdown Current UVLO Start Voltage UVLO Stop Voltage UVLO Start Voltage UVLO Stop Voltage UVLO Start Voltage UVLO Stop Voltage UVLO Hysteresis Shutdown Threshold Hysteresis Internal Bias Voltage (VDD) Oscillator Section Frequency Sync range Sync level Maximum Duty Cycle (each Channel) Minimum Headroom between VOUT Required remote sense amplifier Sync Input Frequency MIC2155 MIC2156 MIC2155 MIC2156 1200 VEN1 VEN2 open open open open VIN1 VIN1 open (each Channel) (each Channel) IVDD -75mA IVDD -50mA 3.97 5.25 0.8V (both O/Ps) (non-switching) Condition Units
2009
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Micrel, Inc. Parameter Minimum On-Time Regulation Feedback Voltage Reference Feedback Bias Current Output Voltage Line Regulation Output Voltage Load Regulation Output Voltage Total Regulation Channel Current Balancing Asynchronous Mode Slave Output Error Amplifier (CH1) Gain Output Sourcing/Sinking Current Error Amplifier (CH2) Gain Transconductance Differential Amplifier Voltage Gain Offset Voltage Output Sourcing Current Range Output Over Voltage Protection Threshold Delay Blanking time Soft Start Internal Soft Start Source Current Current Sense Over Current Trip Point Program Current Comparator Sense Threshold Power Good threshold 88.5 (Senses drop across low-side FET) 1.25 (Latches High) 1.25 4.5V 14.5V; IOUT (VOUT 2.5V) (+/- (+/- VFB=0.7V 14.5 0.08 Condition (each Channel) Note
MIC2155/2156 Units
%Nom
2.75
%Nom
2009
M9999-052709-A (408) 944-0800
Micrel, Inc. Parameter PGOOD Voltage Gate Drivers Rise/Fall Time High Side Drive Resistance Side Drive Resistance Driver Non-Overlap Time (adaptive)
Notes:
MIC2155/2156 Condition IPGOOD Into 3000pF Source Sink Source Sink Source Sink 0.225 Units
Absolute maximum ratings indicate limits beyond which damage component occur. Electrical specifications apply when operating device outside operating ratings. maximum allowable power dissipation function maximum junction temperature, TJ(Max), junction-to-ambient thermal resistance, ambient temperature, maximum allowable power dissipation will result excessive temperature, regulator will into thermal shutdown. device guaranteed function outside operating rating. Specification packaged product only. Minimum on-time before automatic cycle skipping begins. applications section.
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156
Typical Characteristics
VIN1 UVLO Threshold
4.08 UVLO THRESHOLD 4.04 4.02 4.00 3.98 3.96 3.94 3.92 3.90 UVLO Falling UVLO Rising UVLO THRESHOLD 4.06 2.75
VIN2 UVLO Threshold
QUIESCENT CURRENT (mA)
UVLO Rising
Quiescent Current Input Voltage
2.70 2.65 2.60 2.55 2.50 2.45 2.40
UVLO Falling
Switching INPUT VOLTAGE
TEMPERATURE (°C)
TEMPERATURE (°C)
0.10 QUIESCENT CURRENT (mA) 0.09 0.08 0.07 0.06 0.05 0.04 0.03
Quiescent Current Input Voltage
SHUTDOWN CURRENT (mA)
0.20 0.18 0.16 0.14 0.12 0.10 0.08 0.06 0.04 0.02
Shutdown Current Input Voltage
ENABLE1,2
ENABLE THRESHOLD
1.08 1.06 1.04 1.02 1.00 0.98 0.96
Enable Threshold Temperature
Rising Rising
0.02 0.01 Switching INPUT VOLTAGE
INPUT VOLTAGE
TEMPERATURE (°C)
1.00 0.50 FREQUENCY
Change Switching Frequency Input Voltage
4.00 2.00 FREQUENCY 0.00
Change Switching Frequency Temperature
Temperature
Switching
0.00 -0.50 -1.00 -1.50 -2.00 -2.5 INPUT VOLTAGE
-2.00 -4.00 -6.00 -8.00 -10.00 -12.00
TEMPERATURE (°C)
TEMPERATURE (°C)
LOAD (mA)
Load
Switching
Input Voltage
Switching
FEEDBACK VOLTAGE
0.6990 0.6985 0.6980 0.6975 0.6970 0.6965
Feedback Voltage Temperature
INPUT VOLTAGE
TEMPERATURE (°C)
2009
M9999-052709-A (408) 944-0800
0.6960
0.94 Falling 0.92 Falling 0.90 0.88
Micrel, Inc.
MIC2155/2156
Typical Characteristics (cont.)
0.6980 FEEDBACK VOLTAGE 0.6975 0.6970 0.6965 0.6960 0.6955 0.6950 INPUT VOLTAGE
Feedback Voltage Input Voltage
CURRENT (µA)
Current Input Voltage
CURRENT (µA)
Current Temperature
INPUT VOLTAGE
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
RDSON (ohms)
DSON
LowSide Drive
Source/Sink
SINK
SOURCE
TEMPERATURE (°C)
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M9999-052709-A (408) 944-0800
1.022 1.020 1.018 1.016 1.014 1.012 1.010 1.008 1.006 1.004 1.002 1.000
Differential Amplifier Gain Temperature
SOFT START CURRENT (µA)
Soft Start Current Temperature
RDSON (ohms)
DSON
High Side Drive
GAIN
Source/Sink
SINK
SOURCE
Micrel, Inc.
MIC2155/2156
Functional Diagram
MIC2155/6 Block Diagram
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156
Functional Description
MIC2155 MIC2156 two-phase, synchronous buck controllers operating fixed frequency. controllers differ only switching frequency with MIC2155 switching 500kHz phase (1MHz input output) MIC2156 switches 300kHz phase. Some advantages multi-phase operation are: Smaller input output filtering components required because current cancelation higher input output frequency. Faster transient response possible with smaller output filter component values. Load current through each phase half total output current, which allows even heat distribution smaller components. Control circuitry forces better current sharing MOSFETs than paralleling FETs single phase application. controller utilizes voltage-mode control scheme (VMC). Lossless current sharing accomplished sensing voltage across each inductor winding. Lossless overcurrent protection performed sensing voltage across low-side MOSFET on-resistance during off-time. Other features controller are: Overvoltage protection Soft start UVLO Enable Remote sensing Pre-biased output startup Multiple input supplies Power Good signal Frequency synchronization
Figure Startup Sequence
typical output voltage inductor current startup shown Figure
INDUCTOR CURRENT INDUCTOR CURRENT (2A/div) (2A/div)
VOUT (1V/div)
Channel Channel
Time (4ms/div)
Startup typical startup sequence shown Error! Reference source found. (also refer block diagram). enable pins asserted after applied. immediately turned internal releases soft start pin. soft start controls error amplifier voltage. ramps reaches threshold where gate drive enabled MOSFETs start switch very duty cycle. rise soft start voltage controls increase Vout gradually allowing COMP1 voltage rise. 10mV offset current controller keeps Channel low-side drive when output current prevent current from circulating between phases. PGOOD asserted when VOUT reaches PGOOD threshold. 2009
Figure Turn
Soft Start soft start capacitor controls fast output voltage rises controlling COMP risetime. Without soft start fast uncontrolled turn-on requires higher current from input source charge output capacitance. soft start capacitor also controls delay time between enable assertion when VOUT starts rise. Figures show soft start circuitry waveform timing.
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156 disable gate drive, discharge disable VDD. will bring controller into current state. Enable only controls switching Channel Disabling Channel stops switching power FETs Channel which reduces current draw. This improve efficiency when operating output current, especially when large MOSFETs used. Supply Voltages Internal References MIC2155/6 powered from 4.5V 14.5V supply. input supply pins (VIN1 VIN2) connected together most applications. They powered separately configurations with input supply voltages. VIN1 supplies internal LDO, which generates supply voltage. used power gate drive circuitry must externally decoupled power ground pins (PGND1 PGND2). 10µF Ceramic capacitor recommended most applications. AVDD supply Bandgap reference internal analog circuits. small filter (10ohm/0.1µF) connected AVDD recommended help attenuate switching noise from supply. dropout internal regulator causes drop VIN1 below When operating below jumpered VIN1. This bypasses internal prevents from dropping out. simple series pass regulator used limit voltage applications with input voltage that spans above below maximum limit. Figures illustrate examples regulating with external circuitry.
VEN1/2
Slope ISS/CSS
VOUT
Figure Soft Start Waveforms
Figure Soft Start Circuit
output voltage starts rise when approximately diode drop above ground, 0.6V. startup delay output voltage risetime approximated using formula shown below.
Delay 0.6V
10µF
Risetime VOUT soft start discharged under following conditions:
de-asserted UVLO VIN1 pins Overcurrent Overvoltage (latched off)
Figure Regulator
Enable There enable each channels. Asserting will enable Channel gate drive release soft start circuit. De-asserting will 2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156 UVLO Separate UVLO circuits monitor VIN1, VIN2 VDD. Switching Channel inhibited until voltage VIN1 pins greater than their respective UVLO thresholds. gate drive Channel inhibited until VIN2 voltage exceeds UVLO threshold. Individual UVLO thresholds necessary allow proper operation from separate input supplies. VIN1 threshold prevents from switching input voltage properly source voltage. VIN2 UVLO threshold lower than VIN1 allow operation from voltage input. Channel will switch provide regulated output voltage even VIN2 UVLO prevents Channel from switching. Power Good power good signal asserts high when output voltage greater than power good threshold. power good circuit compares portion reference voltage voltage feedback pin. output open drain shown Figure assert high must pulled AVDD through resistor.
AVDD Comparator PGOOD
10µF
Figure Emitter Follower Regulator
internal regulator supply 75mA current drive external MOSFETs. Power dissipation inside MIC2155/6 control divided between power dissipated controller's analog circuitry power dissipated drive circuitry. Drive circuitry power almost always much greater than analog circuitry power. Total regulator power dissipation calculated using following formula:
PDISS VIN1 IIN1 VIN1 Where: total gate charge MOSFETs switching frequency each stage (500kHz MIC2155 300kHz MIC2156) Controller quiescent current (non-switching supply current) some instances, power dissipation inside control limit controller's maximum ambient temperature. example, MIC2155 powered from source driving FETs. each Qg=37nC, total power dissipation MIC2155 PDISS (37nC 500kHz 0.888 maximum operating ambient temperature
BandGap-10%
Figure Power Good
power good signal connected enable other power supplies used sequence other outputs. Oscillator Frequency Synchronization internal oscillator free runs fixed frequency requires external components. oscillator generates clock signals that 180° phase with each other. This forces each channel controller switch 180° phase, which reduces input output ripple current. internal oscillator generates clock signal ramp signal. clock signal terminates switching cycle each channel. ramp voltage Channel compared with output error amplifier regulates output voltage. ramp signal Channel compared with Channel error amplifier output forces output current Channel match Channel
TA(MAX TJ(MAX PDISS TA(MAX 125°C 0.888 TA(MAX 81°C
Using external supply Figure lower power dissipation controller reduce junction temperature supplying externally. Using external regulator, power dissipated controller reduced
PDISS (37nC 500kHz 0.37W Careful selection temperature rise calculations external should done prevent excessively high junction temperature.
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156 minimum value 0.1µF required each bootstrap capacitors, regardless MOSFETs being driven. Larger paralleled MOSFETs require larger capacitance values proper operation. Placement critical. bypass capacitor (CBST) supply pins must located close between pins. etch connections should short, wide direct. ground plane minimize connection impedance recommended. Refer section layout component placement more information. delay between switching MOSFETs necessary prevent both MOSFETs from being same time shorting ground. adaptive gate drive controller monitors switch node (SW1) side driver (LSD1) minimize dead time while preventing both MOSFETs from being same time. This enables broad range MOSFETS without requiring excessive deadtime.
RMP1 CLK1 SYNC RMP2 CLK2 Phase Oscillator
Figure Oscillator Sync Diagram
SYNC input (pin allows MIC2155/6 synchronize external clock signal. When synchronized, each channel switches half synchronization frequency. Limitations synchronization frequency signal amplitude listed electrical characteristics section spec. When used, sync should left open connect). MOSFET Gate Drive Circuitry high-side drive circuit designed switch Nchannel MOSFET. Figure shows diagram gate drive bootstrap circuit. CBST comprise bootstrap circuit, which used supply drive voltage high-side FET. Bootstrap capacitor CBST charged through diode while low-side MOSFET voltage approximately When high-side MOSFET driver turned energy from CBST used charge MOSFET gate, turning FET. MOSFET turns voltage increases approximately VIN. Diode reversed biased CBST pulled high while continuing keep high-side MOSFET high-side drive voltage, which derived from VDD, approximately 4.5V voltage drop across When operating 4.5VIN, without connecting VIN, gate drive voltage high-side could 3.2V. MOSFETs with appropriate threshold should used this situation. voltage bootstrap capacitor drops each time delivers charge turn MOSFET. voltage drop depends gate charge required MOSFET. Most MOSFET specifications specify gate charge voltage. Based this information recommended less than 0.1V, minimum value bootstrap capacitance calculated
QGATE VBST Where: QGATE Total Gate Charge VBST VBST Voltage drop CBST
Figure Gate Drive
dv/dt Induced Turn Low-Side MOSFET high-side MOSFET turns rising dv/dt switch-node forces current through lowside causing glitch FET's gate. Figure illustrates basic mechanism causing this issue. glitch gate greater than FET's turn-on threshold, cause unwanted turn-on lowside while high-side short circuit between input ground would occur that lowers efficiency increases power dissipation both FETs. Additionally, turning low-side during offtime could interfere with overcurrent sensing.
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156 500µA. voltage divider resistors Figure must chosen insure output current amplifier does exceed maximum 500µA.
VOUT VREF where VREF 0.7V 500A gain/phase plot remote sense amplifier Figure shows typical 2MHz bandwidth. Phase 1MHz. R1MAX
(dB) Phase Magnitude 100k FREQUENCY (Hz) -120 -160 -200 PHASE
Figure dv/dt Induced Turn-On
Figure Remote Sense Amplifier Gain/Phase Plot
following steps taken lower gate drive impedance, minimize dv/dt induced current lower FETs susceptibility induced glitch: Chose MOSFET with: high CGS/CGD ratio internal gate resistance resistor between output gate. Insure both gate drive return etch short, inductance connections. 4.5V rated MOSFET because higher gate threshold voltage more immune glitches than 2.5V 3.3V rated FET. Connect supply below RDSON internal driver will lower 4.5V rated MOSFET used. Remote Sense Remote sensing provides accurate output voltage regulation sensing load. Remote sensing makes losses power distribution path. uses unity gain differential amplifier overcome voltage drops both output return (ground) paths. amplifier common mode input range from -0.3V 3.6V. proper remote sense operation, must greater than less than must connected externally supplied with output remote sense amplifier source 2009
typical remote sense configuration shown Figure output remote sense amplifier feeds voltage divider (R1, R4), which connected Channel error amplifier. divider compensation network remote sense same local sense configuration. resistors provide alternate feedback path remote sense connections removed opened. remote sense connections should shorted output voltage will increase close VIN. circuit controller will protect against this type fault since feedback voltage will
+SENSE LOAD -SENSE
Figure Remote Sense
M9999-052709-A (408) 944-0800
Micrel, Inc. Setting Output Voltage Regardless whether remote sensing local output voltage sensing used, output voltage with voltage divider resistors (Figure 12). equation below used calculate Vout.
MIC2155/2156
current during hard short circuits. This helps reduce overall power dissipation converter components during fault.
VOUT VREF Where VREF=0.7V
Current Limit Overcurrent Protection MIC2155/6 uses synchronous (low-side) MOSFETs RDSON sense over current condition. low-side MOSFET used because displays lower parasitic oscillations after switching then upper MOSFET. Additionally, improves accuracy reduces false tripping lower voltage outputs narrow duty cycles since off-time increases duty cycle decreases.
MIC2155/6 200µA
RDSON
Figure Overcurrent Sense Waveforms
Current Limit
Figure Overcurrent Circuit
MIC2155/6 only senses current across side MOSFET Channel since both channels operate parallel. This means total output current limit approximately twice calculated current limit.
Current Limit Setting current limit circuit responds peak inductor current flowing through low-side FET. value estimated with "simple" method more accurately calculated taking inductor ripple current into account. Simple Method Current limit quickly estimated with following equation:
Inductor current flows from lower MOSFET source drain during off-time. drain voltage becomes negative with respect ground inductor current continues flow from Source Drain. This negative voltage proportional instantaneous inductor current times MOSFET RDSON. voltage across lowside becomes even more negative output current increases. overcurrent circuit operates passing known fixed current source (200µA) through resistor RCS. This sets offset voltage (ICS RCS) that compared low-side FET. When (Source Drain current) equal this voltage, MIC2155's over current trigger set, which disables next high side gate drive pulse. After missing high side pulse, over current (OC) trigger reset. next side drive cycle, current still high i.e. another high side pulse missed This effectively reduces overall energy transferred output VOUT starts fall. MIC2155/6 current limit circuit restricts maximum output current. load tries draw additional current output voltage drops until longer within regulation limits. this point (75% nominal output voltage) hiccup current mode initiated protect down stream loads from excessive
IOUT/2 RDSON(MAX)/180µA. Where: RDSON maximum on-resistance side operating junction temperature
Accurate Method designs where ripple current significant when compared IOUT duty cycle operation, calculating current setting resistor should take into account that sensing peak inductor current that there blanking delay approximately 100ns.
2009
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VOUT Efficiency
MIC2155/2156
Figure Overcurrent waveform
equations accurately calculate current limit resistor value shown below:
RIPPLE VOUT IRIPPLE TDLY ISET ISET RDSON(MAX ICS(MIN)
0.3) 3.1A 500kHz 1.5H 16.55 100ns ISET 16.55 16.33 1.5H 16.33 180A Using simple method here would result current limit point lower than expected. This equation sets minimum current limit point converter, maximum will depend actual inductor value resistance MOSFET under current limit conditions. This could region higher should considered ensure that power components within their thermal limits unless thermal protection implemented separately. IRIPPLE
Duty Cycle Switching Frequency Power inductor value TDLY Current limit blanking time 100ns ICS(min) 180µA
Example: Consider 3.3V converter with 1.5µH power inductor efficiency full load. Each channel will supply 500kHz (MIC2155) switching frequency. on-resistance side MOSFET Using simple method
Inductor Current Sensing Current sharing between phases achieved sensing inductor current each phase. Lossless inductor current sensing used, which advantages lower power loss lower cost over using discrete resistor series with inductor. inductor sense circuit shown Figure extracts voltage drop across inductor's winding resistance.
Output Inductor Winding Resistance
180A Using accurate method
Figure Lossless Inductor Current Sense
voltage across capacitor
time constant equal Lo/RL time constant, voltage across capacitor equals Figure plot this equation shows results graphically. assumes inductance 1.5µH,
2009
M9999-052709-A (408) 944-0800
Micrel, Inc. 0.01 (-40dB), C1=0.1µF R1=1.5k. time constants equal diverge same rate. overall impedance, H(s), equals frequencies.
GAIN (dB) H(s)
MIC2155/2156 Channel regulate that voltage. inputs transconductance error amplifier, connected current sense points each channel. error amplifier regulates Channel current monitoring current sense point Channel forcing current sense point Channel equal. offset difference current between channels caused tolerances inductance, DCR, tolerances Additionally, voltage offset cause variations output current sharing. lower currents, these variations force current Channel nominal 10mV offset inhibits Channel low-side MOSFET until output current increases magnitude where voltage across 10mV. This prevents low-side MOSFET Channel from sinking current ground during startup during current operation.
100k FREQUENCY (Hz)
Figure Current Sense Gain/Phase Plot
Current Sharing schematic Figure illustrates current sharing scheme. error amplifier Channel monitors output voltage adjusts duty cycle
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156
Driver VREF
Output1 Inductor Winding Resistance
MIC2155/56
10mV Driver
Output2 Inductor Winding Resistance
Figure Current Sharing Diagram
Startup into Pre-Biased Output Soft start circuitry conventional synchronous buck regulator forces regulator start initially operating minimum duty cycle gradually increasing duty cycle until output voltage reaches regulation. synchronous buck power supply, narrow duty cycle means low-side MOSFET most switching period. output voltage wide time low-side MOSFET discharge output cause high reverse current flow inductor. MIC2155/6 designed turn into pre-biased output without discharging output. Circuitry controller monitors input output voltage forces soft start circuit initially operate proper duty cycle. This allows output turn controlled fashion without discharging output. minimum output voltage proper operation prebias startup circuitry 0.6V. VOUT less than 0.6V, partial discharge VOUT occur.
Separate Input Supplies MIC2155/6 operate from different input supplies with different voltages. Each channels have different input voltage still share current. This allows supply draw power from more than supply. controller will force output current equal. Since output voltage currents channels same, input power drawn from each supply will approximately same. input currents will inversely proportional input voltages each supply. example, total output power efficiency 91%, total input power from both supplies
POUT Each supply contributes approximately half power. PIN1 PIN2 27.5 VIN1 VIN2 3.3V
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
IIN1 IIN2 27.5 27.5
MIC2155/2156 smaller inductor expense higher switching losses slightly lower efficiency. While 300kHz MIC2156 optimized higher efficiency higher output current lower switching frequency requires larger output inductance maintain same peak-topeak output ripple current. peak output ripple current 2-phase converter shown Figure graph shows that peak ripple current function duty cycle. Since each channel 180° phase with other, duty cycle, output ripple currents from each channel cancel output ripple current close zero.
NORMALIZED OUTPUT RIPPLE CURRENT Single Phase
Component Selection, Guidelines Design Example following section outlines procedure designing two-phase synchronous buck converter using MIC2155. This example will following parameters: VOUT 1.8V IOUT Switching frequency (fS) 500kHz/channel (MIC2155) Output Filter Selection output filter comprised output capacitors output inductors. filter designed attenuate output voltage ripple desired value. output filter components also determine well supply responds output current transients. output transients significant, output capacitors should chosen first meet transient specification. output inductor then selected insure filter attenuates output ripple meet specification. second, commonly used method designing filter select inductor value keep ripple current between output current that channel. Then select output capacitance meet output voltage ripple specification output current transient specification. Values inductance, peak currents required choose output inductors. input output voltages inductance value determine peak peak inductor ripple current. Output capacitor selection requires calculation transient current, capacitor current output voltage. There several tradeoffs made when selecting output inductor. Generally, higher inductance values used with higher input voltages. Larger peak peak ripple currents will increase power dissipation inductor MOSFETs. Larger output ripple currents will also require more output capacitance smooth larger ripple current. Smaller peak peak ripple currents require larger inductance value therefore larger more expensive inductor. Higher switching frequencies allow small inductance increase power dissipation inductor core MOSFET switching loss. MIC2155 switches 500kHz/channel designed
Phase
DUTY CYCLE
Figure Phase Output Ripple Current Duty Cycle
this example, with 12V, VOUT 1.8V efficiency 88%, duty cycle
VOUT 0.17 0.88 Figure shows peak-to-peak output ripple current normalized VOUT peak-to-peak output ripple current less than single phase conversion. varies, input voltage that generated highest ripple current should used calculation. this example, assume output transient loading small filter design based output ripple voltage requirement. inductance value calculated equation below.
VOUT VIN(MAX VOUT VIN(MAX IOUT
where: switching frequency ratio ripple current output current VIN(MAX) maximum input voltage
M9999-052709-A (408) 944-0800
2009
Micrel, Inc. IOUT output current each channel total output current converters efficiency this example:
(0.88 0.88 500kHz another inductor value used, ripple current each channel calculated from formula below:
MIC2155/2156 core losses usually insignificant ignored. lower output currents core losses significant contributor. Core loss information usually available from magnetics vendor. this example Cooper HCF1305-1R0 inductor chosen. Core loss this application taken from data sheet 15mW. Winding resistance 1.9mohms Copper loss inductor calculated equation below:
PINDUCTOR(COPPER (IINDUCTOR(RMS) WINDING 15.12 1.9m 0.43
VOUT VIN(MAX VOUT VIN(MAX
(0.88 0.88 500kHz output capacitors less ripple current than each channel because they phase. normalizing factor VOUT 500kHz output ripple current 2-phase configuration approximately: VOUT 0.65 500kHz input output voltage this application, going 2-phase design decreased total output ripple current from 3APP 2.3APP. peak inductor current each channel equal average output current plus half peak peak inductor ripple current. 0.65 IOUT 16.5 inductor current used calculate losses inductor.
IINDUCTOR(RMS) IOUT IOUT
resistance copper wire, RWINDING, increases with temperature. desired, more accurate calculation made maximum ambient temperature temperature rise inductor known. value winding resistance operating temperature calculated with formula below.
WINDING(HOT) WINDING( 0.0042 (TempHOT
Where: TempHOT temperature wire under operating load ambient temperature RWINDING(20) resistance winding room temperature, usually specified manufacturer. this example, approximate power dissipation 0.43W. From manufacturers data sheet this causes 20°C rise inductor temperature. Assuming ambient temperature stayed 20°C, maximum winding resistance would increased from 1.9mohms
WINDING(HOT) 1.9m 0.0042 40°C 20°C) 2.06m
IINDUCTOR(RMS)
15.1A
Maximizing efficiency requires proper selection core material minimizing winding resistance. high frequency operation MIC2155 requires ferrite materials most cost sensitive applications. Lower cost iron powder cores used increase core loss will reduce efficiency power supply. This especially noticeable output power. inductor winding resistance decreases efficiency higher output current levels. winding resistance must minimized although this usually comes expense larger inductor. power dissipated inductor equal core copper losses. higher output loads, 2009
Output Capacitor Selection this example, output capacitors chosen keep output voltage ripple below specified value. output ripple voltage determined capacitors (equivalent series resistance) capacitance. Voltage rating current capability other important factors selecting output capacitor. Ceramic output capacitors most polymer capacitors have very recommended with MIC2155/6. output capacitance usually primary cause output ripple Ceramic very capacitors. minimum value Cout calculated below:
COUT VOPP
Where:
M9999-052709-A (408) 944-0800
Micrel, Inc. VOPP peak peak output voltage ripple peak peak ripple current capacitors channel switching frequency Notice calculation performed switching frequency since capacitors ripple current from both phases. this example, using VOPP 10mV, minimum COUT
10mV 500kHz capacitance value this usually used high current converters because transient output current requirements. this example, 500µF total capacitance used. split into 47µF Ceramic capacitors 150µF Aluminum Polymer capacitors total output ripple combination output capacitance. total ripple calculated below: COUT
MIC2155/2156 inductor following values: 1.0µH, 1.9m Proper sensing voltage across inductor requires RL/L time constant equal time constant.
good range values 0.1µF 1µF. this example chosen 0.22µF. 2.39k 1.9m 0.22F
VOUT [IPP RESR COUT increase reliability, recommended voltage rating capacitor should twice output voltage tantalum greater aluminum electrolytic Ceramic. output capacitor current calculated below: 0.66 power dissipated output capacitors calculated equation below: PDISS(COUT ICOUT(RMS) ICOUT(RMS)
Input Capacitor Selection addition high frequency Ceramic capacitors, larger bulk capacitance, either Ceramic should used help attenuate ripple input supply current input during large output current transients. input capacitors must rated input current power supply. input capacitor current determined maximum output current. graph Figure shows normalized input ripple current duty cycle. Data normalized output current. phase converter operating duty cycle, input current determined from graph:
ICIN IOUT 0.24 7.2A
power dissipated input capacitor
PDISS(CIN) ICIN(RMS) RESR
MORMALIZED INPUT CURRENT DUTY CYCLE Phase Single Phase
RESR
Inductor Current Sense Components circuit values that sense current across inductor calculated once inductor selected. circuit shown Figure
Output Inductor Winding Resistance
Figure Input Current Duty Cycle
Figure Inductor Current Sense
MOSFET Selection External N-Channel logic level power MOSFETs must used high side switches. MOSFET gate source drive voltage MIC2155 regulated internal regulator. Logic level MOSFETs, whose operation specified 4.5V must used. This resistance used calculate losses
2009
M9999-052709-A (408) 944-0800
Micrel, Inc. during MOSFET's conduction time. operating 4.5VIN, without connecting VIN, gate drive voltage high-side could 3.2V. MOSFETs with enhanced gates should used this situation. important note on-resistance MOSFET increases high junction temperature. 75°C rise junction temperature will increase channel resistance MOSFET resistance specified 25°C. This change resistance must accounted when calculating MOSFET power dissipation. Total gate charge charge required turn MOSFET under specified operating conditions (VDS VGS). gate charge supplied MIC2155 gate drive circuit. Gate charge significant source power dissipation controller high switching frequencies generally large MOSFETs that driven. output load this power dissipation noticeable reduction efficiency. average current required drive MOSFETs
where:
MIC2155/2156 power dissipated switching transistor conduction losses during on-time (PCONDUCTION) switching losses that occur during period time when MOSFETs turn (PAC).
PCONDUCTION where:
PCONDUCTION ISWITCH(rms RSWITCH PAC(off PAC(on)
RSWITCH resistance MOSFET switch. Making assumption turn-on turn-off transition times equal, total switching loss
PEAK
Where: switching transition time (typically 15ns 30ns) switching frequency each phase
Current MOSFET Power Dissipation Calculation Under normal operation, high side MOSFET's current greatest when (maximum duty cycle). side MOSFET's current greatest when high (minimum duty cycle). However, MOSFET sees maximum stress during short circuit conditions, where output current equal maximum overcurrent level. calculations below normal operation. calculate stress under short circuit conditions, substitute maximum overcurrent level IOUT(max). value high side switch current
RMS(HIGH SIDE (IOUT(max)
total gate charge high side MOSFETs. This information should obtained from manufacturer's data sheet with VGS. Since current from gate drive comes from input voltage, power dissipated MIC2155 gate drive
PGATE DRIVE
convenient figure merit switching MOSFETs on-resistance times total gate charge (RDSON Qg). Lower numbers translate into higher efficiency. gate charge, logic level MOSFETs good choice with MIC2155. internal that supplies rated 75mA. Exceeding this value could damage regulator cause excessive power dissipation Refer "Supply Voltages Internal Regulator" section this specification additional information. Parameters that important MOSFET switch selection are: Voltage rating resistance Total Gate Charge voltage rating MOSFETs essentially equal input voltage. safety factor should added VDS(max) MOSFETs account voltage spikes circuit parasitics.
RMS(LOW SIDE (IOUT(max)
where: duty cycle converter individual inductor ripple current efficiency converter. Converter efficiency also depends other component parameters that have been selected. design purposes, efficiency estimate 85%-90% used. efficiency more accurately calculated once design complete. assumed efficiency grossly inaccurate, second iteration through design procedure should made.
M9999-052709-A (408) 944-0800
2009
Micrel, Inc. high-side switch, maximum power dissipation
PSWITCH1(DC) RDSON1 1(rms
VDIODE
MIC2155/2156
power dissipated diode
PDIODE where forward voltage peak diode current.
low-side switch, power dissipation
PSWITCH 2(DC) RDSON2 2(rms
switching loss each high-side MOSFETs
(peak
total power dissipation each MOSFET
PFET total PSWITCH1(DC)
External Schottky Diode freewheeling diode parallel with low-side needed keep inductor current flow continuous while both MOSFETs turned (dead time). Dead time necessary prevent current from flowing unimpeded through both MOSFETs. external Schottky diode necessary circuit operation since low-side MOSFET contains parasitic body diode. external diode will improve efficiency lower forward voltage drop compared internal parasitic diode FET. also decrease high frequency noise because schottky diode junction does suffer from reverse recovery. MOSFET body diode used, must rated handle peak average current. body diode have relatively slow reverse recovery time relatively high forward voltage drop. power lost diode proportional forward voltage drop diode. high-side MOSFET starts turn body diode becomes short circuit reverse recovery period, dissipating additional power. diode recovery circuit inductance will cause ringing during high-side MOSFET turn internal diode used, power dissipated during dead time should added PDISS low-side MOSFET. external Schottky diode conducts lower forward voltage preventing body diode MOSFET from turning lower forward voltage drop dissipates less power than body diode. lack reverse recovery mechanism Schottky diode causes less ringing power loss. Depending circuit components operating conditions, external Schottky diode give improvement efficiency. This power dissipation calculated below
ID(ave IOUT
Snubber Design snubber used damp high frequency ringing caused parasitic inductance capacitance buck converter circuit. snubber needed each phases converter. Figure shows simplified schematic buck converter phases. Stray capacitance consists mostly MOSFET's output capacitance (COSS). stray inductance mostly package etch inductance. arrows show resonant current path when high side MOSFET turns This ringing causes stress semiconductors circuit well increased EMI.
COSS1
LSTRAY1
LSTRAY2
LSTRAY3
Sync_buck Controller
COSS2
COUT
LSTRAY4
Figure Output Parasitics
Where dead time when both MOSFETs off. reverse voltage requirement diode
method reducing ringing resistor capacitor lower resonant circuit. circuit Figure shows resistor between switch node ground. Capacitor used block minimize power dissipation resistor. This capacitor value should between times parasitic capacitance MOSFET COSS. capacitor that small will have high impedance prevent resistor from damping ringing. capacitor that large causes unnecessary power dissipation resistor, which lowers efficiency. snubber components should placed close possible low-side MOSFET and/or external schottky diode since contributes most stray capacitance. Placing snubber from using etch that long thin will inductance
M9999-052709-A (408) 944-0800
2009
Micrel, Inc. snubber diminishes effectiveness. proper snubber design requires parasitic inductance capacitance known. method determining these values calculating damping resistor value outlined below. Measure ringing frequency switch node which determined parasitic Define this frequency capacitor (normally least times COSS FET) from switch node ground measure ringing frequency. Define this (lower) frequency solved using values resistor series with generate critical damping. Step First measure ringing frequency switch node voltage when high-side MOSFET turns This ringing characterized equation:
MIC2155/2156
Figure shows snubber circuit damped switch node waveform.
LSTRAY1
LSTRAY2
LSTRAY3
COSS2
LSTRAY4
where
Figure Snubber Circuit
parasitic capacitance inductance Step capacitor, parallel with synchronous MOSFET, capacitor value should approximately times COSS Measure frequency switch node ringing,
snubber capacitor, charged discharged each switching cycle. energy stored dissipated snubber resistor, times switching period. This power calculated equation below.
PSNUBBER VIN2 where: switching frequency each phase input voltage
Compensation Output Voltage Loop voltage regulation, filter power stage section shown Figure error amplifier Channel used regulate output voltage compensate voltage regulation loop. voltage output that designed type (PID) compensation. Type compensation compensating zeros, poles pole origin. figure also shows transfer function each section. Compensation necessary insure control loop adequate bandwidth phase margin properly respond input voltage output current transients. High gain frequencies needed accurate output voltage regulation. Attenuation near switching frequency prevents switching frequency noise from interfering with control loop.
Define
Combining equations derive parasitic capacitance
solved re-arranging equation
f1)2 Step Calculate damping resistor. Critical damping occurs
Solving
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
Driver VREF 0.7V RESR Modulator
MIC2155/2156
Filter
Figure Voltage Loop Transfer Functions
this analysis, filter phase design combined into one. inductances parallel output capacitance total Cout. parallel combination ESRs. output load represented resistor output filter contains complex double pole formed capacitor inductor zero from output capacitor it's ESR. transfer function filter
Gfilter(s)
VOUT
modulator, filter voltage divider gains multiplied together show open loop gain these parts.
GFILTER GMOD
where:
This transfer function plotted Figure frequency, transfer function gain equals modulator gain times voltage divider gain. frequency increases toward filter resonant frequency, gain starts peak. increase gain's amplitude equals Just above resonant frequency, gain drops -40db/decade rate. phase quickly drops from almost 180° before phase boost zero brings back -90°. Higher values will cause phase drop quickly. well damped, system phase will change more slowly.
GAIN/PHASE -135 -180 100k FREQUENCY Phase Gain
Modulator Gain proportional input voltage inversely proportional internal ramp voltage generated oscillator. MIC2155/6 peak-peak ramp voltage
GMOD RAMP
output voltage divider attenuates VOUT feeds back error amplifier. divider gain
Figure Transfer Function
frequency approaches zero frequency (Fz), 2009
M9999-052709-A (408) 944-0800
Micrel, Inc. formed it's ESR, slope gain curve changes from -40db/dec. -20db/dec phase increases. zero causes phase boost. Ceramic capacitors, with their smaller values capacitance ESR, push zero phase boost higher frequencies, which allow phase from filter drop closer -180°. system will close being unstable overall open loop gain crosses while phase close -180°. output capacitance and/or high, zero moves lower frequency helps boost phase, leading more stable system. error amplifier type which zeros, poles pole origin. This type error amplifier works well when Ceramic output capacitors make majority COUT because introduces extra zero that helps improve phase margin.
Gea(s)
MIC2155/2156
Figure Type Error Amplifier Gain/Phase
where:
Figure shows bode plot error amplifier transfer function.
Error Amplifier Design Procedure
Step Decide crossover frequency maximize transient response, open loop bandwidth should made reasonably high. Initially, bandwidth selected 1/10 output switching frequency. This improved once design built measurements made. initial bandwidth 100kHz 2155 60kHz 2156 good choices. Step Determine gain required crossover frequency GBoost much gain boost needed open loop transfer function crosses pre-determined crossover frequency. This measured plotting Gvd(s) transfer function estimated with following formula:
GBoost
Where: filter resonant frequency open loop bandwidth chosen Step zero formed Cout voltage divider attenuation Vm=amplitude internal sawtooth ramp (Vm=1) Vin= Input voltage power supply
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
Step Determine gain boost needed crossover frequency (fc) Typically, phase margin used most applications. This good tradeoff between overdamped system (slower response transients) underdamped system (overshoot unstable response transients). also allows some margin component tolerances variations ambient temperature changes. phase margin crossover frequency (fc) determined plotting Gvd(s) phase bode plot estimated with following formula:
MIC2155/2156
proper phase margin. high, will provide attenuation switching frequency, which could lead jitter switching waveform instability under certain conditions.
additional phase boost required from error amplifier
Boost
Step Determine frequencies frequencies zero pole (fz2 fp1) calculated desired amount phase boost crossover frequency (fc).
sin[Boost sin[Boost sin[Boost sin[Boost
Calculating Error Amplifier Component Values Once pole zero frequencies have been fixed, error amplifier's resistor capacitor values calculated. This value chosen first. other component values calculated from value suggested. chosen high, very large high impedances could sensitive noise. remote sense amplifier used, must large enough than more than 500µA current drawn from amplifier. value determined from mid-band gain error amplifier. This gain depends frequencies poles, zeros filter resonant frequency. Based amount gain necessary crossover frequency mid-band gain value calculated using following formula.
other component values calculated follows:
Step Determine frequency frequency zero, fz1, initially one-fifth resonant frequency. low, will force frequency gain impact transient response. high, will enough phase boost resonant frequency. This could cause conditional stability, which when phase drops below 180° before gain crosses 0dB. gain should drop this situation, this lead unstable system.
Step Determine frequency This high frequency pole, which useful additional attenuation switching frequency. should initially half switching frequency. low, will lower phase margin crossover frequency, making difficult achieve
Compensation Current Sharing Loop control circuitry Channel forces channel's output current match current Channel Channel error amplifier compares inductor currents channels adjusts duty cycle Channel control output current. block diagram shown Figure
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156
Channel Transconductance
Modulator
Filter
Ramp
Driver
Channel
Figure Current Sharing Loop Transfer Functions
applied voltage Unlike voltage output amplifier used Channel compensation, transconductance amplifier used Channel compensation since only pole/zero combination required compensation. transconductance amplifier transfer function
Gea(s)
VOUT GFILTER( open loop transfer function GMOD GFILTER( Cz1) (VIN VOUT Cz1)
GAIN/PHASE
where: external components connected COMP2 transconductance internal amplifier. pole zero frequencies are: fPOLE fZERO gain modulator
where peak-to-peak amplitude internal sawtooth. gain feedback circuit output current divide gMOD
loop inherently stable because phase shift only degrees. error amplifier pole zero selected achieve desired crossover frequency. this example, desired crossover frequency 50kHz. transfer function filter, modulator feedback plotted Figure
VOUT 1.8V 1.5µH Gain
Phase -100 100k FREQUENCY
filter transfer function output current over
Figure Current Sharing Loop Gain/Phase
2009
M9999-052709-A (408) 944-0800
Micrel, Inc. gain boost required 50kHz 28dB which gain gain frequencies above zero
GMID
MIC2155/2156 Figure shows layout example that minimizes inductance.
typical 1.25mS, solving
GMID 1.25mS
zero frequency crossover frequency.
800pF compensated open loop gain/phase plot shown Figure
LSFET
Load
HSFET
Figure Layout
134.17466 GAIN/PHASE bb(f) fZERO 10kHz 50kHz Phase margin Gain
-100 -179.9424
Phase
-200
100k FREQUENCY
Figure Compensated Current Sharing Loop Gain/Phase
General Layout Component Placement There three basic types currents switching power supply high di/dt, moderate di/dt Examples each shown Figure
Figure Current Diagram
buck converter, high di/dt currents 0.5A/ns range generated MOSFETs switching off. These fast switching currents flow high lowside MOSFETs, external freewheeling schottky diode input capacitor. Fast switching currents also flow gate drive return etch between controller power FETs. that switching speed 10nH piece etch generates across itself. Therefore, attention proper layout techniques essential. Traces that have high di/dt currents must kept short wide. Additionally power ground plane should used adjacent layer help minimize etch inductance.
Moderate di/dt currents flow inductor output capacitor. Although layout critical, still important minimize inductance using short, wide traces ground plane. Figure shows etch connecting inductor output shaped force current flow past output capacitor before reaching output terminal output load). This minimizes series inductance between inductor capacitor, which improves ability capacitor filter ripple. Additionally, inductor current large component requires wide trace minimize voltage drop power dissipation. currents high current buck converter require wide etch paths minimize voltage drop power dissipation. input output current mainly near maximum output power, inductor current also predominately requires ample etch reduce copper loss, reduce temperature rise improve efficiency. Minimizing voltage drops output ground path helps improve output voltage regulation configurations without remote voltage sensing. gate drive connections both high-side low-side MOSFETs must each have their return current path. high-side MOSFET's source connected switch node returns back controller's pin. high-side gate drive return (switch node) traces should routed each other adjacent layers minimize inductance. These traces swing between ground should routed away from voltage noise sensitive analog etch components. low-side MOSFET return path power ground. High di/dt currents flow low-side gate drive return paths. These must kept away from noise sensitive signal traces signal
M9999-052709-A (408) 944-0800
2009
Micrel, Inc. ground planes. Ceramic capacitors recommended most decoupling filtering applications because their impedance small size. Depending application, most dielectrics (X5R, X7R, NPO) acceptable, however, type Ceramic capacitor dielectrics recommended their large change capacitance over temperature voltage.
Design Layout Checklist
MIC2155/2156 These analog signals should referenced decoupled analog ground plane: AVDD, SYNC, PGOOD, COMP1, COMP2, FB2, EA2, VOUT, FB1, AGND Place current sharing components (that connect across inductor) related filtering components close FB2, EA2+ VOUT pins (18, 20). traces connecting inductors these components should routed close together minimize pickup radiation. Place overcurrent sense resistor close (pin trace coming from switch node this resistor high dv/dt should routed away from other noise sensitive components traces. remote sense traces must routed close together adjacent layers minimize noise pickup. traces should routed away from switch node, inductors, MOSFETs other high dv/dt di/dt sources.
Ceramic capacitor placed between drain LSFET source. MOSFET gate drive traces must inductance routed away from noise sensitive analog signals, components ground planes. signal power ground planes must separated prevent high current fast switching signals from interfering with level, noise sensitive analog signals. These planes should connected only point, next MIC2155/6 controller. following signals their components should decoupled referenced power ground plane: VIN1, VIN2, VDD, PGND1, PGND2
2009
M9999-052709-A (408) 944-0800
Micrel, Inc.
MIC2155/2156
Package Information
32-Pin MLF® (ML)
MICREL, INC. 2180 FORTUNE DRIVE JOSE, 95131
(408) 944-0800 (408) 474-1000 http://www.micrel.com
information furnished Micrel this data sheet believed accurate reliable. However, responsibility assumed Micrel use. Micrel reserves right change circuitry specifications time without notification customer. Micrel Products designed authorized components life support appliances, devices systems where malfunction product reasonably expected result personal injury. Life support devices systems devices systems that intended surgical implant into body support sustain life, whose failure perform reasonably expected result significant injury user. Purchaser's sale Micrel Products life support appliances, devices systems Purchaser's risk Purchaser agrees fully indemnify Micrel damages resulting from such sale. 2009 Micrel, Incorporated.
2009
M9999-052709-A (408) 944-0800

2-Phase, Single Output, Synchronous Buck Control MIC2155 MIC2156

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