LTC3810 synchronous step-down switching regulator controller that dire
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LTC3810 100V Current Mode Synchronous Switching Regulator Controller DESCRIPTION
LTC3810 synchronous step-down switching regulator controller that directly step-down voltages from 100V, making ideal telecom automotive applications. LTC3810 uses constant on-time valley current control architecture deliver very duty cycles with accurate cycle-by-cycle current limit, without requiring sense resistor. precise internal reference provides 0.5% accuracy. high bandwidth (25MHz) error amplifier provides very fast line load transient response. Large gate drivers allow LTC3810 drive multiple MOSFETs higher current applications. operating frequency selected external resistor compensated variations also synchronized external clock switching-noise sensitive applications. shutdown allows LTC3810 turned off, reducing supply current 240A. Integrated bias control generates gate drive power from input supply during start-up when output shortcircuit occurs, with addition small external SOT23 MOSFET. When regulation, power derived from output higher efficiency.
High Voltage Operation: 100V Large Gate Drivers Current Sense Resistor Required Dual N-Channel MOSFET Synchronous Drive Extremely Fast Transient Response ±0.5% 0.8V Voltage Reference Programmable Output Voltage Tracking/Soft-Start Generates Driver Supply from Input Supply Synchronizable External Clock Selectable Pulse Skip Mode Operation Power Good Output Voltage Monitor Adjustable On-Time/Frequency: tON(MIN) 100ns Adjustable Cycle-by-Cycle Current Limit Programmable Undervoltage Lockout Output Overvoltage Protection 28-Pin SSOP Package
APPLICATIONS
Telecom Base Station Power Supplies Networking Equipment, Servers Automotive Industrial Control Systems
Lare registered trademarks Linear Technology Corporation. other trademarks property their respective owners. Protected U.S. Patents including 5481178, 5847554, 6304066, 6476589, 6580258, 6677210, 6774611.
TYPICAL APPLICATION
High Efficiency High Voltage Step-Down Converter
261k PGOOD VRNG 1000pF LTC3810 0.1F EXTVCC DRVCC INTVCC SENSE+ SGND SENSE- BGRTN Si7456DP RFB2 MBR1100 RFB1 Si7456DP VOUT 12V/6A NDRV BOOST 100k ZXMN10A07F
Efficiency Load Current
EFFICIENCY
100V
MODE/SYNC SS/TRACK SHDN
200k
COUT 270F
LOAD
3810 TA01b
47pF
3810 TA01
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LTC3810 ABSOLUTE MAXIMUM RATINGS
(Note
CONFIGURATION
VIEW VRNG PGOOD MODE/SYNC BOOST SENSE+ SENSE- BGRTN DRVCC INTVCC EXTVCC NDRV
Supply Voltages INTVCC, DRVCC -0.3V (DRVCC BGRTN), (BOOST -0.3V BOOST -0.3V 114V BGRTN EXTVCC -0.3V (NDRV INTVCC) Voltage -0.3V SENSE+ Voltage 100V Voltage -0.3V 100V SS/TRACK Voltage -0.3V PGOOD Voltage -0.3V VRNG, VON, MODE/SYNC, SHDN, UVIN Voltages. -0.3V PLL/LPF Voltages. -0.3V 2.7V INTVCC, EXTVCC Currents .50mA Operating Temperature Range (Note LTC3810E. -40°C 85°C LTC3810I. -40°C 125°C Junction Temperature (Notes 125°C Storage Temperature Range. -65°C 150°C Lead Temperature (Soldering, sec) 300°C
PLL/LPF SS/TRACK SGND SHDN UVIN
PACKAGE 28-LEAD PLASTIC SSOP TJMAX 125°C, 100°C/W
ORDER INFORMATION
LEAD FREE FINISH LTC3810EG#PBF LTC3810IG#PBF LEAD BASED FINISH LTC3810EG LTC3810IG TAPE REEL LTC3810EG#TRPBF LTC3810IG#TRPBF TAPE REEL LTC3810EG#TR LTC3810IG#TR PART MARKING LTC3810EG LTC3810IG PART MARKING LTC3810EG LTC3810IG PACKAGE DESCRIPTION 28-Lead Plastic SSOP 28-Lead Plastic SSOP PACKAGE DESCRIPTION 28-Lead Plastic SSOP 28-Lead Plastic SSOP TEMPERATURE RANGE -40°C 85°C -40°C 125°C TEMPERATURE RANGE -40°C 85°C -40°C 125°C
Consult Marketing parts specified with wider operating temperature ranges. more information lead free part marking, http://www.linear.com/leadfree/ more information tape reel specifications,
denotes specifications which apply over full operating temperature range, otherwise specifications 25°C, INTVCC DRVCC VBOOST VRNG SHDN UVIN VEXTVCC VNDRV 10V, VMODE/SYNC VSENSE+ VSENSE VBGRTN =0V, unless otherwise specified.
SYMBOL Main Control Loop INTVCC PARAMETER INTVCC Supply Voltage INTVCC Supply Current INTVCC Shutdown Current CONDITIONS
ELECTRICAL CHARACTERISTICS
6.35
UNITS
SHDN 1.5V, INTVCC 9.5V (Notes SHDN
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LTC3810 ELECTRICAL CHARACTERISTICS
SYMBOL IBOOST PARAMETER BOOST Supply Current Feedback Voltage
denotes specifications which apply over full operating temperature range, otherwise specifications 25°C, INTVCC DRVCC VBOOST VRNG SHDN UVIN VEXTVCC VNDRV 10V, VMODE/SYNC VSENSE+ VSENSE VBGRTN unless otherwise specified.
CONDITIONS SHDN 1.5V (Note SHDN (Note 85°C -40°C 85°C -40°C 125°C (I-Grade) INTVCC (Note VRNG 0.76V VRNG 0.76V VRNG INTVCC, 0.76V VRNG 0.84V VRNG 0.84V VRNG INTVCC, 0.84V 0.8V (Note VMODE/SYNC Rising MODE/SYNC 0.75 Rising Falling Hysteresis INTVCC Rising, INDRV 100A INTVCC Rising, NDRV INTVCC EXTVCC INTVCC Rising, NDRV INTVCC, EXTVCC INTVCC Falling 100A 300A 2000A
0.796 0.794 0.792 0.792
VFB,LINE VSENSE(MAX)
Feedback Voltage Line Regulation Maximum Current Sense Threshold
0.800 0.800 0.800 0.800 0.002 -300 -200 0.88 0.80 0.10 1.85
0.804 0.806 0.806 0.808 0.02
UNITS
VSENSE(MIN)
Minimum Current Sense Threshold
IVFB AVOL(EA) VMODE/SYNC IMODE/SYNC VSHDN ISHDN VVINUV
Feedback Current Error Amplifier Open Loop Gain Error Unity-Gain Crossover Frequency MODE/SYNC Threshold MODE/SYNC Current Shutdown Threshold SHDN Input Current Undervoltage Lockout
0.86 0.78 0.07 6.05 6.05 11.7
0.85 0.92 0.82 0.12 6.35 6.35 12.3
VVCCUV
INTVCC Undervoltage Lockout Linear Regulator Mode External Supply Mode Trickle-Charge Mode
Oscillator Phase-Locked Loop On-Time tON(MIN) tOFF(MIN) tON(PLL) Minimum On-Time Minimum Off-Time Modulation Range Down Modulation Modulation Phase Detector Output Current Sinking Capability Sourcing Capability Driver Peak Source Current Driver Pull-Down RDS(ON) Driver Peak Source Current Driver Pull-Down RDS(ON) PGOOD Upper Threshold PGOOD Lower Threshold PGOOD Hysteresis
1.55
2.15
100A, VPLL/LPF 0.6V 100A, VPLL/LPF 1.8V fPLLIN fPLLIN
IPLL/LPF
Driver IBG,PEAK RBG,SINK ITG,PEAK RTG,SINK PGOOD Output VFBOV VFB,HYST
12.5 -12.5
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Rising Falling Returning
-7.5
LTC3810 ELECTRICAL CHARACTERISTICS
SYMBOL VPGOOD IPGOOD Delay Tracking ISS/TRACK VFB,TRACK Regulators VEXTVCC PARAMETER PGOOD Voltage PGOOD Leakage Current PGOOD Delay SS/TRACK Source Current Feedback Voltage Tracking
denotes specifications which apply over full operating temperature range, otherwise specifications 25°C, INTVCC DRVCC VBOOST VRNG SHDN UVIN VEXTVCC VNDRV 10V, VMODE/SYNC VSENSE+ VSENSE- VBGRTN unless otherwise specified.
CONDITIONS IPGOOD VPGOOD Falling VSS/TRACK 0.5V VTRACK 1.2V (Note VTRACK 0.5V, 1.2V (Note -0.018 UNITS
0.48
0.52
VINTVCC,1 VEXTVCC,1 VLOADREG,1 VINTVCC,2 VLOADREG,2 INDRV INDRVTO VCCSR ICCSR
EXTVCC Switchover Voltage EXTVCC Rising EXTVCC Hysteresis INTVCC Voltage from EXTVCC VEXTVCC VINTVCC Dropout INTVCC Load Regulation from EXTVCC INTVCC Voltage from NDRV Regulator INTVCC Load Regulation from NDRV Current into NDRV Linear Regulator Timeout Enable Threshold Maximum Supply Voltage Maximum Current into NDRV/INTVCC
10.5V VEXTVCC 20mA, VEXTVCC 9.1V 20mA, VEXTVCC Linear Regulator Operation 20mA, VEXTVCC VNDRV VINTVCC
0.01 0.01
10.6 10.6
Trickle Charger Shunt Regulator Trickle Charger Shunt Regulator, INTVCC 16.7V (Note
Note Stresses beyond those listed under Absolute Maximum Ratings cause permanent damage device. Exposure Absolute Maximum Rating condition extended periods affect device reliability lifetime. Note LTC3810E guaranteed meet performance specifications from 85°C. Specifications over -40°C 85°C operating temperature range assured design, characterization correlation with statistical process controls. LTC3810I guaranteed meet perfomance specifications over full -40°C 125°C operating temperature range. Note calculated from ambient temperature power dissipation according following formula: LTC3810: 100°C/W) PARAMETER Maximum MOSFET Gate Drive INTVCC INTVCC LTC3810 100V 6.35V 6.2V
Note LTC3810 tested feedback loop that servos reference voltage with forced voltage between Note dynamic input supply current higher power MOSFET gate charging being delivered switching frequency fOSC). Note Guaranteed design. subject test. Note This includes overtemperature protection that intended protect device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection active. Continuous operation above specified maximum operating junction temperature impair device reliability. Note current into NDRV INTVCC.
LTC3810-5 4.5V 4.2V
LTC3812-5 4.5V 4.2V
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LTC3810 TYPICAL PERFORMANCE CHARACTERISTICS
Load Transient Response
VOUT 100mV/ INTVCC 5V/DIV VOUT 5V/DIV 50V/DIV 5A/DIV 50s/DIV LOAD STEP FRONT PAGE CIRCUIT
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Start-Up
VOUT 10V/DIV SS/TRACK 4V/DIV 5A/DIV
3810
Short-Circuit/ Fault Timeout Operation
INTVCC
IOUT 5A/DIV
500s/DIV ILOAD MODE/SYNC FRONT PAGE CIRCUIT
10ms/DIV RSHORT FRONT PAGE CIRCUIT
3810
Short-Circuit/ Foldback Operation
VOUT 5V/DIV 0.5V/ VOUT 5V/DIV SS/TRACK 0.5V/DIV 0.5V/DIV 5A/DIV 200s/DIV FRONT PAGE CIRCUIT
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Tracking
VOUT 100mV/ 0.5V/DIV 2A/DIV
Pulse Skip Mode Operation
SS/TRACK
5A/DIV
500s/DIV ILOAD MODE/SYNC FRONT PAGE CIRCUIT
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20s/DIV IOUT 100mA MODE/SYNC INTVCC FRONT PAGE CIRCUIT
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Efficiency Input Voltage
IOUT EFFICIENCY EFFICIENCY IOUT 0.5A 250kHz FRONT PAGE CIRCUIT INPUT VOLTAGE
Efficiency Load Current
FREQUENCY (kHz)
Frequency Input Voltage
IOUT
IOUT
VOUT Si7850 MOSFETs MODE/SYNC INTVCC 250kHz LOAD CURRENT
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MODE/SYNC FRONT PAGE CIRCUIT INPUT VOLTAGE
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LTC3810 TYPICAL PERFORMANCE CHARACTERISTICS
Frequency Load Current
FREQUENCY (kHz) PULSE SKIP FRONT PAGE CIRCUIT
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Current Sense Threshold Voltage
VRNG 10000 CURRENT SENSE THRESHOLD (mV) -100 -200 -300 -400 VOLTAGE
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On-Time Current
INTVCC
FORCED CONTINUOUS
1.4V ON-TIME (ns) 0.7V 0.5V 1000
1000 CURRENT 10000
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LOAD CURRENT
On-Time Voltage
ON-TIME (ns) ON-TIME (ns) VOLTAGE
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On-Time Temperature
300A MAXIMUM CURRENT SENSE THRESHOLD (mV)
Current Limit Foldback
VRNG INTVCC
300A
TEMPERATURE (°C)
38125
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Maximum Current Sense Threshold VRNG Voltage
MAXIMUM CURRENT SENSE THRESHOLD (mV) MAXIMUM CURRENT SENSE THRESHOLD (mV)
Maximum Current Sense Threshold Temperature
0.803 VRNG INTVCC 0.802 REFERENCE VOLTAGE 0.801 0.800 0.799 0.798
Feedback Reference Voltage Temperature
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VRNG VOLTAGE
TEMPERATURE (°C)
0.797
TEMPERATURE (°C)
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3810
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LTC3810 TYPICAL PERFORMANCE CHARACTERISTICS
Driver Peak Source Current Temperature
1.50 VBOOST VINTVCC 1.25 PEAK SOURCE CURRENT RDS(ON) 1.00 0.75 0.50 0.25 PEAK SOURCE CURRENT TEMPERATURE (°C)
Driver Pull-Down RDS(ON) Temperature
VBOOST VINTVCC
Driver Peak Source Current Supply Voltage
TEMPERATURE (°C)
DRVCC/BOOST VOLTAGE
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Driver Pull-Down RDS(ON) Supply Voltage
RESISTANCE
EXTVCC Resistance Dropout Temperature
INTVCC Current Temperature
INTVCC CURRENT (mA) TEMPERATURE (°C)
RDS(ON)
DRVCC/BOOST VOLTAGE
TEMPERATURE (°C)
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3810
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INTVCC Shutdown Current Temperature
INTVCC CURRENT (mA) TEMPERATURE (°C) INTVCC CURRENT
INTVCC Current INTVCC Voltage
INTVCC VOLTAGE
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3810
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LTC3810 TYPICAL PERFORMANCE CHARACTERISTICS
INTVCC Shutdown Current INTVCC Voltage
SS/TRACK CURRENT INTVCC CURRENT INTVCC VOLTAGE
SS/TRACK Pull-Up Current Temperature
TEMPERATURE (°C)
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Voltage Load Current
VOLTAGE LOAD CURRENT
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Shutdown Threshold Temperature
SHUTDOWN THRESHOLD TEMPERATURE (°C)
VRNG FRONT PAGE CIRCUIT
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LTC3810 FUNCTIONS
(Pin On-Time Current Input. resistor from this one-shot timer current thereby switching frequency. (Pin On-Time Voltage Input. Voltage trip point on-time comparator. Tying this output voltage external resistive divider from output makes on-time proportional VOUT. comparator defaults 0.7V when grounded defaults 2.4V when connected INTVCC. this INTVCC high VOUT applications lower value. VRNG (Pin Sense Voltage Limit Set. voltage this sets nominal sense voltage maximum output current from 0.5V resistive divider from INTVCC. nominal sense voltage defaults 95mV when this tied ground, 215mV when tied INTVCC. PGOOD (Pin Power Good Output. Open-drain logic output that pulled ground when output voltage between ±10% regulation point. output voltage must regulation least 120s before power good output pulled ground. MODE/SYNC (Pin Pulse Skip Mode Enable/Sync Pin. This multifunction provides pulse skip mode enable/ disable control external clock input phase detector. Pulling this below 0.8V external logic-level synchronization signal disables pulse skip mode operation forces continuous operation. Pulling this above 0.8V enables pulse skip mode operation. clock input, phase-locked loop will force rising gate signal synchronized with rising edge clock signal.This also connected feedback resistor divider from secondary winding inductor regulate second output voltage. (Pin Error Amplifier Compensation Point Current Control Threshold. current comparator threshold increases with control voltage. voltage ranges from 2.6V with 1.2V corresponding zero sense voltage (zero current). (Pin Feedback Input. Connect through resistor divider network VOUT output voltage. PLL/LPF (Pin 10): phase-locked loop's lowpass filter tied this pin. voltage this defaults 1.2V when synchronized with external clock MODE/SYNC pin. SS/TRACK (Pin 11): Soft-Start/Tracking Input. softstart, capacitor ground this sets ramp rate output voltage (approximately 0.6s/F). coincident ratiometric tracking, connect this resistive divider between voltage tracked ground. SGND (Pin 12): Signal Ground. small-signal components should connect this ground eventually connect PGND point. SHDN (Pin 13): Shutdown Pin. Pulling this below 1.5V will shut down LTC3810, turn both external MOSFET switches reduce quiescent supply current 240A. UVIN (Pin 14): UVLO Input. This input internal UVLO compared internal 0.8V reference. external resistor divider connected this input supply program undervoltage lockout voltage. When UVIN less than 0.8V, LTC3810 shut down. NDRV (Pin 15): Drive Output External Pass Device Linear Regulator INTVCC. Connect gate external NMOS pass device pull-up resistor input voltage VIN. EXTVCC (Pin 16): External Driver Supply Voltage. When this voltage exceeds 6.7V, internal switch connects this INTVCC through turns external MOSFET connected NDRV, that controller gate drive drawn from EXTVCC. INTVCC (Pin 17): Main Supply Pin. internal circuits except output drivers powered from this pin. INTVCC should bypassed ground (Pin with least 0.1F capacitor close proximity LTC3810. DRVCC (Pin 18): Driver Supply Pin. DRVCC supplies power output driver. This normally connected INTVCC. DRVCC should bypassed BGRTN (Pin with (X5R better) capacitor close proximity LTC3810.
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LTC3810 FUNCTIONS
(Pin 19): Bottom Gate Drive. drives gate bottom N-channel synchronous switch MOSFET. This swings from BGRTN DRVCC. BGRTN (Pin 20): Bottom Gate Return. This connects source pull-down MOSFET driver normally connected ground. Connecting negative supply this allows synchronous MOSFET gate pulled below ground help prevent false turn-on during high dV/dt transitions node. Applications Information section more details. SENSE+, SENSE- (Pin 21): Current Sense Comparator Input. input current comparator normally connected unless using sense resistor. input used accurately kelvin sense bottom side sense resistor MOSFET. (Pin 26): Switch Node Connection Inductor Bootstrap Capacitor. Voltage swing this from Schottky diode (external) voltage drop below ground VIN. (Pin 27): Gate Drive. drives gate N-channel synchronous switch MOSFET. driver draws power from BOOST returns pin, providing true floating drive MOSFET. BOOST (Pin 28): Gate Driver Supply. BOOST supplies power floating driver. BOOST should bypassed with (X5R better) 0.1F capacitor. additional fast recovery Schottky diode from DRVCC BOOST will create complete floating charge-pumped supply BOOST.
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LTC3810 FUNCTIONAL DIAGRAM
INTVCC EXTVCC NDRV INTVCC RUV1 RUV2 UVIN 0.8V INTVCC MODE LOGIC
NDRV
INTVCC
6.2V
200A
INTVCC EXTVCC
0.8V MODE/SYNC PLL/LPF
270A
PLL-SYNC
1.4A
100nA TIMEOUT LOGIC FCNT VVON (76pF) IION
6.7V BOOST SENSE+ DRVCC
ICMP
IREV
SWITCH LOGIC
VOUT
SHDN
BGRTN CVCC
COUT
1.4V VRNG ITH' FOLDBACK 2.6V 0.7V OVERTEMP SENSE
SENSE- PGOOD
RFB1
0.72V
FAULT SHDN
SGND 0.88V
RFB2
0.8V
1.5V SS/TRACK SHDN
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LTC3810 OPERATION
Main Control Loop LTC3810 current mode controller DC/DC stepdown converters. normal operation, MOSFET turned fixed interval determined one-shot timer (OST). When MOSFET turned off, bottom MOSFET turned until current comparator ICMP trips, restarting one-shot timer initiating next cycle. Inductor current determined sensing voltage between SENSE- SENSE+ pins using sense resistor bottom MOSFET on-resistance. voltage sets comparator threshold corresponding inductor valley current. fast 25MHz error amplifier adjusts this voltage comparing feedback signal internal 0.8V reference voltage. load current increases, causes drop feedback voltage relative reference. voltage then rises until average inductor current again matches load current. operating frequency determined implicitly MOSFET on-time duty cycle required maintain regulation. one-shot timer generates time that proportional ideal duty cycle, thus holding frequency approximately constant with changes VIN. nominal frequency adjusted with external resistor RON. applications with stringent constant frequency requirements, LTC3810 synchronized with external clock. programming nominal frequency same external clock frequency, LTC3810
PULSE SKIP MODE FORCED CONTINUOUS
behaves constant frequency part against load supply variations. Pulling SHDN forces controller into shutdown state, turning both Forcing voltage above 1.5V will turn device. Pulse Skip Mode LTC3810 operate modes selectable with MODE/SYNC pin-pulse skip mode forced continuous mode (see Figure Pulse skip mode selected when increased efficiency light loads desired (see Figure this mode, bottom MOSFET turned when inductor current reverses minimize efficiency loss reverse current flow gate charge switching. load currents, will drop below zero current level (1.2V) shutting both switches. Both switches will remain with output capacitor supplying load current until voltage rises above zero current level initiate another cycle. this mode, frequency proportional load current light loads. Pulse skip mode operation disabled comparator when MODE/SYNC brought below 0.8V, forcing continuous synchronous operation. Forced continuous mode less efficient resistive losses, advantage better transient response currents, approximately constant frequency operation, ability maintain regulation when sinking current.
EFFICIENCY
DECREASING LOAD CURRENT
PULSE SKIP MODE FORCED CONTINUOUS LOAD
3810
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0.01
Figure Comparison Inductor Current Waveforms Pulse Skip Mode Forced Continuous Operation
Figure Efficiency Pulse Skip Mode Forced Continuous Mode
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LTC3810 OPERATION
Fault Monitoring/Protection Constant on-time current mode architecture provides accurate cycle-by-cycle current limit protection-a feature that very important protecting high voltage power supply from output short circuits. cycle-by-cycle current monitor guarantees that inductor current will never exceed value programmed VRNG pin. Foldback current limiting provides further protection output shorted ground. drops, buffered current threshold voltage ITHB pulled down clamped This reduces inductor valley current level one-sixth maximum value approaches Foldback current limiting disabled start-up. Overvoltage undervoltage comparators pull PGOOD output output feedback voltage exits ±10% window around regulation point after internal 120s power mask timer expires. Furthermore, overvoltage condition, turned turned immediately held until overvoltage condition clears. LTC3810 provides undervoltage lockout comparators-one INTVCC/DRVCC supply input supply VIN. INTVCC threshold 6.2V guarantee that MOSFETs have sufficient gate drive voltage before turning threshold (UVIN pin) 0.8V with hysteresis which allows programming threshold with appropriate resistor divider connected VIN. either comparator inputs under threshold, LTC3810 shut down drivers turned off. Strong Gate Drivers LTC3810 contains very impedance drivers capable supplying amps current slew large MOSFET gates quickly. This minimizes transition losses allows paralleling MOSFETs higher current applications. 100V floating high side driver drives side MOSFET side driver drives bottom side MOSFET (see Figure bottom side driver supplied directly from DRVCC pin. MOSFET drivers biased from floating bootstrap capacitor, which normally recharged during each cycle through external diode from DRVCC when MOSFET turns off. pulse skip mode operation, where possible that bottom MOSFET will extended period time, internal timeout guarantees that bottom MOSFET turned least once every on-time period refresh bootstrap capacitor. bottom driver additional feature that helps minimize possibility external MOSFET shoot-through. When MOSFET turns switch node dV/dt pulls bottom MOSFET's internal gate through Miller capacitance, even when bottom driver holding gate terminal ground. gate pulled high enough, shoot-through between side bottom side MOSFETs occur. prevent this from occurring, bottom driver return brought separate (BGRTN) that negative supply used reduce effect Miller pull-up. example, supply used BGRTN, switch node dV/dt could pull gate before bottom MOSFET more than across
DRVCC
LTC3810
DRVCC
BOOST
VOUT
BGRTN
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COUT
Figure Floating Driver Supply Negative Return
IC/Driver Supply Power LTC3810's internal control circuitry bottom MOSFET drivers operate from supply voltage (INTVCC, DRVCC pins) range 6.2V 14V. LTC3810 integrated linear regulator controllers easily generate this IC/driver supply from either high voltage input from output voltage. best efficiency supply derived from input voltage during start-up then derived from lower voltage output soon output higher than 6.7V. Alternatively, supply derived from input continuously output
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LTC3810 OPERATION
6.7V external supply appropriate range used. LTC3810 will automatically detect which mode being used operate properly. four possible operating modes generating this supply summarized follows (see Figure LTC3810 generates start-up supply from small external SOT23 N-channel MOSFET acting linear regulator with drain connected gate controlled LTC3810's internal linear regulator controller through NDRV pin. soon output voltage reaches 6.7V, IC/driver supply derived from output through internal dropout regulator optimize efficiency. output lost short, LTC3810 goes through repeated duty cycle soft-start cycles (with drivers shut between) attempt bring output without burning SOT23 MOSFET. This scheme eliminates long start-up times associated with conventional trickle charger using external MOSFET quickly charge IC/driver supply capacitors (CINTVCC, CDRVCC). Similar except that external MOSFET used continuous IC/driver power instead just
Mode MOSFET Start-Up Only
270A 270A
start-up. MOSFET sized proper dissipation driver shutdown/restart VOUT 6.7V disabled. This scheme less efficient necessary VOUT 6.7V boost network desired. Trickle charge mode provides even simpler approach eliminating external MOSFET. IC/driver supply capacitors charged through single high valued resistor connected input supply. When INTVCC voltage reaches turn-on threshold (automatically raised from 6.7V provide extra headroom start-up), drivers turn begin charging output capacitor. When output reaches 6.7V, IC/driver power derived from output. trickle-charge mode, supply capacitors must have sufficient capacitance such that they discharged below INTVCC threshold before output high enough take over else power supply will start. voltage supply available. simplest approach voltage supply (between 6.2V 14V) available connected directly IC/driver supply pins.
Mode MOSFET Continuous
NDRV INTVCC LTC3810
NDRV
INTVCC LTC3810
EXTVCC
VOUT (>6.7V)
EXTVCC
Mode Trickle Charge Mode
Mode External Supply
NDRV INTVCC LTC3810
NDRV
INTVCC LTC3810
6.2V
3810
EXTVCC
VOUT
EXTVCC
Figure Operating Modes IC/Driver Supply
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LTC3810 APPLICATIONS INFORMATION
basic LTC3810 application circuit shown first page this data sheet. External component selection primarily determined maximum input voltage load current begins with selection sense resistance power MOSFET switches. LTC3810 uses either sense resistor on-resistance synchronous power MOSFET determining inductor current. desired amount ripple current operating frequency largely determines inductor value. Next, selected ability handle large current into converter COUT chosen with enough meet output voltage ripple transient specification. Finally, loop compensation components selected meet required transient/phase margin specifications. Maximum Sense Voltage VRNG Inductor current determined measuring voltage across sense resistance that appears between SENSE- SENSE+ pins. maximum sense voltage voltage applied VRNG equal approximately: VSENSE(MAX) 0.173VRNG 0.026 current mode control loop will allow inductor current valleys exceed VSENSE(MAX)/RSENSE. practice, should allow some margin variations LTC3810 external component values good guide selecting sense resistance
RSENSE VSENSE(MAX)
sense resistor. Using sense resistor provides well defined current limit, adds cost reduces efficiency. Alternatively, eliminate sense resistor bottom MOSFET current sense element simply connecting SENSE+ lower MOSFET drain SENSE MOSFET source. This improves efficiency, must carefully choose MOSFET on-resistance, discussed below. Power MOSFET Selection LTC3810 requires external N-channel power MOSFETs, (main) switch bottom (synchronous) switch. Important parameters power MOSFETs breakdown voltage BVDSS, threshold voltage V(GS)TH, on-resistance RDS(ON), input capacitance maximum current IDS(MAX). When bottom MOSFET used current sense element, particular attention must paid on-resistance. MOSFET on-resistance typically specified with maximum value RDS(ON)(MAX) 25°C. this case, additional margin required accommodate rise MOSFET on-resistance with temperature: RDS(ON)(MAX) RSENSE
term normalization factor (unity 25°C) accounting significant variation on-resistance with temperature (see Figure typically varies from 0.4%/°C 1.0%/°C depending particular MOSFET used.
NORMALIZED ON-RESISTANCE
external resistive divider from INTVCC used voltage VRNG between 0.5V resulting nominal sense voltages 60mV 320mV. Additionally, VRNG tied SGND INTVCC which case nominal sense voltage defaults 95mV 215mV, respectively. Connecting SENSE+ SENSE- Pins LTC3810 used with without sense resistor. When using sense resistor, place between source bottom MOSFET, PGND. Connect SENSE+ SENSE- pins bottom
JUNCTION TEMPERATURE (°C)
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Figure RDS(ON) Temperature
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LTC3810 APPLICATIONS INFORMATION
most important parameter high voltage applications breakdown voltage BVDSS. Both bottom MOSFETs will full input voltage plus additional ringing switch node across drain-to-source during off-time must chosen with appropriate breakdown specification. Since most MOSFETs 100V range have higher thresholds (typically VGS(MIN) 6V), LTC3810 designed used with 6.2V gate drive supply (DRVCC pin). maximum efficiency, on-resistance RDS(ON) input capacitance should minimized. RDS(ON) minimizes conduction losses input capacitance minimizes transition losses. MOSFET input capacitance combination several components taken from typical "gate charge" curve included most data sheets (Figure
MILLER EFFECT CMILLER QA)/VDS
MOSFET directly specified MOSFET data sheets. CRSS specified sometimes definitions these parameters included. When controller operating continuous mode duty cycles bottom MOSFETs given MainSwitchDutyCycle VOUT VOUT
SynchronousSwitchDutyCycle
power dissipation main synchronous MOSFETs maximum output current given PTOP VOUT (IMAX )RDS(ON) VIN2 (RDR )(CMILLER VTH(IL) VTH(IL) PBOT VOUT (IMAX
)RDS(0N)
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Figure Gate Charge Characteristic
curve generated forcing constant input current into gate common source, current source loaded stage then plotting gate voltage versus time. initial slope effect gate-to-source gate-to-drain capacitance. flat portion curve result Miller multiplication effect drain-to-gate capacitance drain drops voltage across current source load. upper sloping line drain-to-gate accumulation capacitance gate-to-source capacitance. Miller charge (the increase coulombs horizontal axis from while curve flat) specified given drain voltage, adjusted different voltages multiplying ratio application curve specified values. estimate CMILLER term take change gate charge from points manufacturers data sheet divide stated voltage specified. CMILLER most important selection criteria determining transition loss term
where temperature dependency RDS(ON), effective driver resistance (approximately VMILLER), drain potential change drain potential particular application. VTH(IL) data sheet specified typical gate threshold voltage specified power MOSFET data sheet specified drain current. CMILLER calculated capacitance using gate charge curve from MOSFET data sheet technique described above. Both MOSFETs have losses while topside N-channel equation incudes additional term transition losses, which peak highest input voltage. high input voltage duty cycle applications that typical LTC3810, transition losses dominate loss term therefore using higher RDS(ON) device with lower CMILLER usually provides highest efficiency. synchronous MOSFET losses greatest high input voltage when switch duty factor during short circuit when synchronous switch close 100%
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LTC3810 APPLICATIONS INFORMATION
period. Since there transition loss term synchronous MOSFET, optimal efficiency obtained minimizing RDS(ON)- using larger MOSFETs paralleling multiple MOSFETs. Multiple MOSFETs used parallel lower RDS(ON) meet current thermal requirements desired. LTC3810 contains large impedance drivers capable driving large gate capacitances without significantly slowing transition times. fact, when driving MOSFETs with very gate charge, sometimes helpful slow down drivers adding small gate resistors less) reduce noise caused fast transitions. Operating Frequency choice operating frequency tradeoff between efficiency component size. frequency operation improves efficiency reducing MOSFET switching losses requires larger inductance and/or capacitance order maintain output ripple voltage. operating frequency LTC3810 applications determined implicitly one-shot timer that controls on-time, tON, MOSFET switch. on-time current voltage according VVON (76pF) IION
1000 VOUT SWITCHING FREQUENCY (kHz) SWITCHING FREQUENCY (kHz) VOUT 3.3V VOUT 2.5V VOUT
Tying resistor from yields on-time inversely proportional VIN. step-down converter, this results approximately constant frequency operation input supply varies: VOUT VVON RON(76pF)
hold frequency constant during output voltage changes, VOUT resistive divider from VOUT when VOUT 2.4V. internal clamps that limit input one-shot timer. tied below 0.7V, input one-shot clamped 0.7V. Similarly, tied above 2.4V, input clamped 2.4V. high VOUT applications, INTVCC. Figures show relates switching frequency several common output voltages. Changes load current magnitude will cause frequency shift. Parasitic resistance MOSFET switches inductor reduce effective voltage across inductance, resulting increased duty cycle load current increases. lengthening on-time slightly current increases, constant frequency operation maintained. This accomplished with resistive divider from VOUT. values required will depend parasitic resistances specific
1000
VOUT 1.5V
VOUT 3.3V
VOUT
1000
3810 F07a
1000
3810 F07b
Figure Switching Frequency (VON
Figure Switching Frequency (VON INTVCC)
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LTC3810 APPLICATIONS INFORMATION
application. good starting point feed about voltage change shown Figure Place capacitance filter variations switching frequency.
INTVCC RVON1 200k RVON2 CVON 0.01F 100k LTC3810
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Inductor Selection Given desired input output voltages, inductor value operating frequency determine ripple current: VOUT VOUT
Figure Correcting Frequency Shift with Load Current Changes
Minimum Off-Time Dropout Operation minimum off-time, tOFF(MIN), smallest amount time that LTC3810 capable turning bottom MOSFET, tripping current comparator turning MOSFET back off. This time generally about 250ns. minimum off-time limit imposes maximum duty cycle tON/(tON tOFF(MIN)). maximum duty cycle reached, dropping input voltage example, then output will drop regulation. minimum input voltage avoid dropout VIN(MIN) VOUT tOFF(MIN)
Lower ripple current reduces core losses inductor, losses output capacitors output voltage ripple. Highest efficiency operation obtained frequency with small ripple current. However, achieving this requires large inductor. There tradeoff between component size, efficiency operating frequency. reasonable starting point choose ripple current that about IOUT(MAX). largest ripple current occurs highest VIN. guarantee that ripple current does exceed specified maximum, inductance should chosen according VOUT IL(MAX) VOUT VIN(MAX)
plot maximum duty cycle frequency shown Figure
SWITCHING FREQUENCY (MHz)
Once value known, type inductor must selected. High efficiency converters generally cannot afford core loss found cost powdered iron cores, forcing more expensive ferrite, molypermalloy Kool cores. variety inductors designed high current, voltage applications available from manufacturers such Sumida, Panasonic, Coiltronics, Coilcraft Toko. Schottky Diode Selection Schottky diode shown front page schematic conducts during dead time between conduction power MOSFET switches. intended prevent body diode bottom MOSFET from turning storing charge during dead time, which cause modest (about efficiency loss. diode rated about half fifth full load current since only fraction duty cycle. order
DROPOUT REGION
0.25 0.50 0.75 DUTY CYCLE (VOUT/VIN)
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Figure Maximum Switching Frequency Duty Cycle
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LTC3810 APPLICATIONS INFORMATION
diode effective, inductance between bottom MOSFET must small possible, mandating that these components placed adjacently. diode omitted efficiency loss tolerable. Input Capacitor Selection continuous mode, drain current MOSFET approximately square wave duty cycle VOUT/VIN which must supplied input capacitor. prevent large input transients, input capacitor sized maximum current given VOUT ICIN(RMS) IO(MAX) VOUT This formula maximum 2VOUT, where IRMS IO(MAX)/2. This simple worst-case condition commonly used design because even significant deviations offer much relief. Note that ripple current ratings from capacitor manufacturers often based only 2000 hours life. This makes advisable further derate capacitor choose capacitor rated higher temperature than required. Several capacitors also placed parallel meet size height requirements design. Because tantalum OS-CON capacitors available voltages above 30V, ceramics aluminum electrolytics must used regulators with input supplies above 30V. Ceramic capacitors have advantage very handle high current, ceramics with high voltage ratings 50V) available with more than microfarads capacitance. Furthermore, ceramics have high voltage coefficients which means that capacitance values decrease even more when used rated voltage. type ceramics recommended their lower voltage temperature coefficients. Another consideration when using ceramics their high which, properly damped, result excessive voltage stress power MOSFETs. Aluminum electrolytics have much higher bulk capacitance, they have higher lower current ratings. good approach combination aluminum electrolytics bulk capacitance ceramics current. current cannot handled aluminum capacitors alone, when used together, percentage current that will supplied aluminum capacitor reduced approximately: IRMS,ALUM 100% (8fCRESR where RESR aluminum capacitor overall capacitance ceramic capacitors. Using aluminum electrolytic with ceramic also helps damp high ceramic, minimizing ringing. Output Capacitor Selection selection COUT primarily determined required minimize voltage ripple. output ripple (VOUT) approximately equal VOUT 8fCOUT
Since increases with input voltage, output ripple highest maximum input voltage. also significant effect load transient response. Fast load transitions output will appear voltage across COUT until feedback loop LTC3810 change inductor current match load current value. Typically, once requirement satisfied capacitance adequate filtering required current rating. Manufacturers such Nichicon, Nippon Chemi-Con Sanyo should considered high performance throughhole capacitors. OS-CON (organic semiconductor dielectric) capacitor available from Sanyo lowest product size aluminum electrolytic somewhat higher price. additional ceramic capacitor parallel with OS-CON capacitors recommended reduce effect their lead inductance. surface mount applications, multiple capacitors placed parallel required meet ESR, current handling load step requirements. tantalum, special polymer aluminum electrolytic capacitors available surface mount packages. Special polymer capacitors offer very have lower capacitance density than other types. Tantalum capacitors have highest capacitance density important only types
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LTC3810 APPLICATIONS INFORMATION
that have been surge tested switching power supplies. Several excellent surge-tested choices AVX, TPSV KEMET T510 series. Aluminum electrolytic capacitors have significantly higher ESR, used cost-driven applications providing that consideration given ripple current ratings longterm reliability. Other capacitor types include Panasonic Sanyo POSCAPs. Output Voltage LTC3810 output voltage resistor divider according following formula: VOUT 0.8V RFB1 RFB2 MOSFET Driver Supply (CB, external bootstrap capacitor, connected BOOST supplies gate drive voltage topside MOSFET. This capacitor charged through diode from DRVCC when switch node low. When MOSFET turns switch node rises BOOST rises approximately INTVCC. boost capacitor needs store about times gate charge required MOSFET. most applications 0.1F 0.47F dielectric capacitor adequate. reverse breakdown external diode, must greater than VIN(MAX). Another important consideration external diode reverse recovery reverse leakage, either which cause excessive reverse current flow full reverse voltage. reverse current times reverse voltage exceeds maximum allowable power dissipation, diode damaged. best results, ultrafast recovery diode such MMDL770T1. Bottom MOSFET Driver Return Supply (BGRTN) bottom gate driver, switches from DRVCC BGRTN where BGRTN voltage between ground -5V. just keep simple always connect BGRTN ground? high voltage switching converters, switch node dV/dt many volts/ns, which will pull gate bottom MOSFET through Miller capacitance. this Miller current, times internal gate resistance MOSFET plus driver resistance, exceeds threshold FET, shoot-through will occur. using negative supply BGRTN, pulled below ground when turning bottom MOSFET off. This provides extra volts margin before gate reaches turn-on threshold MOSFET. aware that maximum voltage difference between DRVCC BGRTN 14V. example, VBGRTN -2V, maximum voltage DRVCC instead 14V. IC/MOSFET Driver Supplies (INTVCC DRVCC) LTC3810 drivers supplied from DRVCC BOOST pins (see Figure which have absolute maximum voltage 14V. Since main supply voltage,
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external resistor divider connected output shown Functional Diagram, allowing remote voltage sensing. resultant feedback signal compared with internal precision 800mV voltage reference error amplifier. internal reference guaranteed tolerance Tolerance feedback resistors will additional error output voltage. 0.1% resistors recommended. Input Voltage Undervoltage Lockout resistor divider connected from input supply UVIN (see Functional Diagram) used program input supply undervoltage lockout thresholds. When rising voltage UVIN reaches 0.88V, LTC3810 turns when falling voltage UVIN drops below 0.8V, LTC3810 shut down-providing hysteresis. input voltage UVLO thresholds resistor divider according following formulas: VIN,FALLING 0.8V VIN,RISING 0.88V RUV1 RUV2 RUV1 RUV2
input supply undervoltage lockout needed, disabled connecting UVIN INTVCC
LTC3810 APPLICATIONS INFORMATION
typically much higher than separate supply power (INTVCC) driver power (DRVCC) must used. LTC3810 integrated bias supply control circuitry that allows IC/driver supply easily generated from and/or VOUT with minimal external components. There four ways this shown simplified schematics Figure explained following sections. Using Linear Regulator INTVCC/DRVCC Supply Mode small external SOT23 MOSFET, controlled NDRV pin, used generate start-up supply from VIN. small SOT23 package used because NMOS continuously only during brief start-up period. soon output voltage reaches 6.7V, LTC3810 turns external NMOS LTC3810 regulates supply from EXTVCC (connected VOUT VOUT derived boost network) through internal dropout regulator. this mode work properly, EXTVCC must range 6.7V EXTVCC 15V. VOUT 6.7V, charge pump extra winding used raise EXTVCC proper voltage, alternatively, Mode should used explained later this section. VOUT shorted otherwise goes below minimum 6.5V threshold, MOSFET connected turned back maintain supply. However output cannot brought within timeout period, drivers turned prevent SOT23 MOSFET
FAULT TIMEOUT ENABLED SS/TRACK
from overheating. Soft-start cycles then attempted duty cycle intervals bring output back (see Figure 10). This fault timeout operation enabled choosing choosing RNDRV such that resistor current INDRV greater than 270A using following formulas: RNDRV PMOSFET(MAX) 270A
where QG(TOP) QG(BOTTOM)
threshold voltage MOSFET. value NDRV also affects IN(MIN) follows: VIN(MIN) VINTVCC(MIN) (40A) RNDRV where VINTVCC(MIN) normally driving 100V MOSFETs. minimum enough, consider reducing RNDRV and/or using darlington instead NMOS reduce ~1.4V. When using RNDRV equal computed value, LTC3810 will enable duty cycle soft-start retries only when desired maximum power dissipation, PMOSFET(MAX), MOSFET exceeded leave drivers continuously otherwise. shutoff/restart times function TRACK/SS capacitor value.
DRIVER THRESHOLD DRIVER POWER FROM VOUT DRIVER POWER FROM EXTVCC THRESHOLD
ISS/TRACK (SOURCE) ISS/TRACK (SINK)
DRIVER POWER FROM START-UP
VOUT SHORT-CIRCUIT EVENT START-UP INTO SHORT-CIRCUIT
TG/BG
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Figure Fault Timeout Operation
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LTC3810 APPLICATIONS INFORMATION
external NMOS linear regulator should standard threshold type (i.e., logic level threshold). rate charge INTVCC from controlled LTC3810 approximately regardless size capacitor connected INTVCC pin. charging current this capacitor approximately: INTVCC required RNDRV value results unacceptable value VIN(MIN) (see Equation fault timeout operation also disabled connecting 500k resistor from SS/TRACK INTVCC. Using Trickle Charge Mode Trickle charge mode selected shorting NDRV INTVCC connecting EXTVCC VOUT. Trickle charge mode advantage requiring external MOSFET takes longer start slow charge CINTVCC CDRVCC through RPULLUP (tDELAY 0.77 RPULLUP CDRVCC) usually requires larger INTVCC/ DRVCC capacitor values hold supply voltage during start-up. Once INTVCC/DRVCC voltage reaches trickle charge threshold 12V, drivers will turn start discharging CINTVCC/CDRVCC rate determined driver current order ensure proper startup, CINTVCC/CDRVCC must chosen large enough that EXTVCC voltage reaches switchover threshold 6.7V before CINTVCC/CDRVCC discharges below falling threshold This ensured CINTVCC CDRVCC COUT arger IMAX VOUT(REG) where gate drive current (f)(QG(TOP) QG(BOTTOM)) IMAX maximum inductor current selected VRNG. RPULLUP value should fall following range ensure proper start-up: RPULLUP (VIN(MAX)-14V)/ICCSR RPULLUP (VIN(MIN)-12V)/IQ,SHUTDOWN
safe operating area (SOA) external NMOS should chosen that capacitor charging does damage NMOS. Excessive values capacitor unnecessary should avoided. Typically values work well. more design requirement this mode minimum soft-start capacitor value. fault timeout enabled when SS/TRACK voltage greater than This gives power supply time bring output before starts timeout sequence. prevent timeout sequence from starting prematurely during start-up, minimum value necessary ensure that VSS/TRACK until VEXTVCC 6.7V. ensure this, choose: COUT (2.3 10-6)/IOUT(MAX) Mode should used VOUT outside 6.7V EXTVCC operating range extra complexity charge pump extra inductor winding wanted boost this voltage above 6.7V. this mode, EXTVCC grounded NMOS chosen handle worstcase power dissipation:
PMOSFET VIN(MAX)
f)(QG(TOP) QG(BOTTOM)
operate properly, fault timeout operation must disabled choosing RNDRV (VIN(MAX) VTH)/270A
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LTC3810 APPLICATIONS INFORMATION
Using External Supply Connected INTVCC/ DRVCC Pins external supply available between 6.2V 14V, supply connected directly INTVCC/DRVCC pins. this mode, INTVCC, EXTVCC NDRV must shorted together. INTVCC/DRVCC Supply EXTVCC Connection LTC3810 contains internal dropout regulator produce INTVCC/DRVCC supply from EXTVCC voltage. This regulator turns when EXTVCC above 6.7V remains until EXTVCC drops below 6.4V. This allows IC/MOSFET power derived from output output derived boost network during normal operation from external NMOS from during start-up short-circuit. Using EXTVCC this results significant efficiency gains compared what would possible when deriving this power continuously from typically much higher voltage. EXTVCC connection also allows power supply configured trickle charge mode which starts with high valued "bleed" resistor connected from INTVCC charge INTVCC capacitor. soon output rises above 6.7V internal EXTVCC regulator takes over before INTVCC capacitor discharges below threshold. When EXTVCC regulator active, EXTVCC supply 50mA RMS. apply more than EXTVCC pin. following list summarizes possible connections EXTVCC: EXTVCC grounded. This connection will require INTVCC powered continuously from external NMOS from resulting efficiency penalty high high input voltages. EXTVCC connected directly VOUT. This normal connection 6.7V VOUT provides highest efficiency. power supply will start using external NMOS bleed resistor until output supply available. EXTVCC connected output-derived boost network. VOUT 6.7V. voltage output boosted using charge pump flyback winding greater than 6.7V. EXTVCC connected INTVCC. This required connection EXTVCC INTVCC connected external supply where external supply 6.2V VEXT 15V. Applications using large MOSFETs with high input voltage high frequency operation result large EXTVCC current. Therefore, good design practice verify that maximum junction temperature rating current rating within maximum limits. Typically, most EXTVCC current consists MOSFET gates current. continuous mode operation, this EXTVCC current IEXTVCC QG(TOP) QG(BOTTOM) 50mA junction temperature estimated from equations given Note Electrical Characteristics follows: IEXTVCC (VEXTVCC VINTVCC)(100°C/W) 125°C absolute maximum ratings exceeded, consider using external supply connected directly INTVCC pin.
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LTC3810 APPLICATIONS INFORMATION
FEEDBACK LOOP/COMPENSATION Feedback Loop Types typical LTC3810 circuit, feedback loop consists modulator, output filter load, feedback amplifier with compensation network. these components affect loop behavior must accounted loop compensation. modulator output filter consists internal current comparator, output MOSFET drivers external MOSFETs, inductor output capacitor. Current mode control eliminates effect inductor moving inner loop, reducing first order system. From feedback loop point view, looks like linear voltage controlled current source from VOUT gain equal (IMAXROUT)/1.2V. fairly benign behavior typical loop compensation frequencies with significant phase shift appearing half switching frequency. external output capacitor load cause first order roll output ROUTCOUT pole frequency, with attendant phase shift. This roll what filters waveform, resulting desired output voltage. output capacitor also contributes zero COUTRESR frequency which adds back phase cancels first order roll off. far, response loop pretty well user's control. modulator fundamental piece LTC3810 design external output capacitor usually chosen based regulation load current requirements without considering loop response. feedback amplifier, other hand, gives handle with which adjust response. goal have 180° phase shift loop regulates), something less than 360° phase shift (preferably about 300°) point that loop gain falls 0dB, i.e., crossover frequency, with much gain possible frequencies below crossover frequency. Since modulator/output filter first order system with maximum phase shift frequencies below fSW/4) feedback amplifier adds another phase shift, some phase boost required crossover frequency achieve good phase margin. zero below crossover frequency, this zero provide enough phase boost achieve desired phase margin only requirement compensation will guarantee that gain below zero frequencies above fSW/4. zero above crossover frequency, feedback amplifier will probably required provide phase boost. most LTC3810 applications, Type compensation will provide enough phase boost; however some applications where high bandwidth required with ceramics lots bulk capacitance, Type compensation necessary provide additional phase boost. types compensation networks, "Type "Type shown Figures When component values chosen properly, these networks provide "phase bump" crossover frequency. Type uses single pole-zero pair provide about phase boost while Type uses poles zeros provide 150° phase boost.
VREF GAIN (dB) PHASE (DEG)
-6dB/OCT GAIN -6dB/OCT
FREQ PHASE -180 -270 -360
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Figure Type Schematic Transfer Function
GAIN (dB)
PHASE (DEG)
-6dB/OCT
GAIN
+6dB/OCT
-6dB/OCT FREQ
VREF
PHASE
-180 -270 -360
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Figure Type Schematic Transfer Function
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LTC3810 APPLICATIONS INFORMATION
Feedback Component Selection Selecting values typical Type Type loop nontrivial task. applications shown this data sheet show typical values, optimized power components shown. They should give acceptable performance with similar power components, even major power component changed significantly. Applications that require optimized transient response will require recalculation compensation values specifically circuit question. underlying mathematics complex, component values calculated straightforward manner know gain phase modulator crossover frequency. Modulator gain phase obtained three ways: measured directly from breadboard, appropriate parasitic values known, simulated generated from modulator transfer function. Measurement will give more accurate results, simulation transfer function often close enough give working system. measure modulator gain phase directly, wire breadboard with LTC3810 actual MOSFETs, inductor input output capacitors that final design will use. This breadboard should appropriate construction techniques high speed analog circuitry: bypass capacitors located close LTC3810, long wires connecting components, appropriately sized ground returns, etc. Wire feedback amplifier with 0.1F feedback capacitor from 100k resistor from VOUT Choose bias resistor (RB) required desired output voltage. Disconnect from ground connect signal generator source output network analyzer inject test signal into loop. Measure gain phase from output node positive terminal output capacitor. Make sure analyzer's input coupled that voltages present both VOUT nodes don't corrupt measurements damage analyzer. breadboard measurement practical, SPICE simulation used generate approximate gain/ phase curves. Plug expected capacitor, inductor MOSFET values into following SPICE deck generate plot VOUT VITH with gain phase degrees. Refer your SPICE manual details generate this plot.
*3810 modulator gain/phase *2006 Linear Technology *this file simulates simplified model *the LTC3810 generating v(out)/v(ith) *bode plot .param rdson=.0135 ;MOSFET rdson .param Vrng=2 ;use INTVCC .param .param .param ground vsnsmax={0.173*Vrng-0.026} Imax={vsnsmax/rdson} DL=4 ;inductor ripple current
*inductor current *output cout out2 270u ;capacitor value resr out2 0.018 ;capacitor *load Rout load resistor vstim stimulus 10meg .probe .end
Mathematical software such MATHCAD MATLAB also used generate plots using following transfer function modulator: H(s) VSENSE(MAX) RDS(ON) RESR COUT COUT
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LTC3810 APPLICATIONS INFORMATION
With gain/phase plot hand, loop crossover frequency chosen. Usually curves look something like Figure Choose crossover frequency about switching frequency maximum bandwidth. Although tempting beyond fSW/4, remember that significant phase shift occurs half switching frequency that isn't modeled above H(s) equation PSPICE code. Note gain (GAIN, phase (PHASE, degrees) this point. desired feedback amplifier gain will -GAIN make loop gain this frequency. calculate needed phase boost, assuming target phase margin: BOOST (PHASE 30°) required BOOST less than 60°, Type loop used successfully, saving external components. BOOST values greater than usually require Type loops satisfactory performance. Finally, choose convenient resistor value (10k usually good value). calculate remaining values: constant used calculations) chosen crossover frequency 10(GAIN/20) (this converts GAIN absolute gain) TYPE Loop: BOOST
PHASE (DEG)
GAIN (dB)
GAIN
PHASE
-180 FREQUENCY (Hz)
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Figure Transfer Function Buck Modulator
TYPE Loop: BOOST tan2 (R1) VOUT VREF SPICE mathematical software used generate gain/phase plots compensated power supply sanity check component values before trying them actual hardware. software, following transfer function: T(s) A(s)H(s)
(R1) VOUT VREF
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LTC3810 APPLICATIONS INFORMATION
where H(s) given Equation A(s) depends compensation circuit used: Type RFB1 Pulse Skip Mode Operation MODE/SYNC MODE/SYNC determines whether bottom MOSFET remains when current reverses inductor. Tying this above 0.8V threshold enables pulse skip mode operation where bottom MOSFET turns when inductor current reverses. load current which current reverses discontinuous operation begins depends amplitude inductor ripple current will vary with changes VIN. Tying MODE/SYNC below 0.8V threshold forces continuous synchronous operation, allowing current reverse light loads maintaining high frequency operation. prevent forcing current back into main power supply, potentially boosting input supply dangerous voltage level, forced continuous mode operation disabled when TRACK/SS voltage below reference voltage during soft-start tracking. During these periods, PGOOD signal forced low. addition providing logic input force continuous operation, MODE/SYNC provides mean maintain flyback winding output when primary operating pulse skip mode. secondary output VOUT2 normally shown Figure turns ratio transformer. However, controller goes into pulse skip mode halts switching light primary load current, then VOUT2 will droop. external resistor divider from VOUT2 MODE/SYNC sets minimum
Type
(R1+
SPICE, replace VSTIM line previous PSPICE code with following code generate gain/phase plot V(out)/V(outin):
rfb1 outin 52.5k rfb2 eithx ithx laplace {0.8-v(vfb)} {1/(1+s/1000)} eith 210k outin ;delete this line Type 120p ;delete this line Type vstim outin dc=0 ac=1m
LTC3810 SGND PGND
1N4148
VOUT2 COUT2 VOUT1 COUT
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Figure Secondary Output Loop
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LTC3810 APPLICATIONS INFORMATION
voltage VOUT2(MIN) below which continuous operation forced until VOUT2 risen above minimum. VOUT2(MIN) 0.8V
Table
MODE/SYNC Voltage: 0.75V Voltage: 0.85V Feedback Resistors Ext. Clock CONDITION Forced Continuous Current Reversal Enabled Pulse Skip Mode Operation Current Reversal Regulating Secondary Winding Forced Continuous Current Reversal Enabled
lies same percentage below typical value maximum lies above Consult MOSFET manufacturer further guidelines. further limit current event short-circuit ground, LTC3810 includes foldback current limiting. output falls more than 60%, then maximum sense voltage progressively lowered about tenth full value. aware also that when fault timeout enabled external NMOS regulator, over current limit cause output fall below minimum 6.5V threshold. This condition will cause linear regulator timeout/restart sequence described Linear Regulator Timeout section this condition persists. Soft-Start Tracking LTC3810 ability either soft-start itself with capacitor track output another supply. When device configured soft-start itself, capacitor should connected TRACK/SS pin. LTC3810 quiescent current shutdown state ~240A) SHDN voltage below 1.5V. TRACK/SS actively pulled ground this shutdown state. Once SHDN voltage above 1.5V, LTC3810 powered soft-start current 1.4A then starts charge soft-start capacitor CSS. Note that soft-start achieved limiting maximum output current controller controlling ramp rate output voltage. Current foldback disabled during this soft-start phase. During soft-start phase, LTC3810 ramping reference voltage until reaches 0.8V. force continuous mode also disabled PGOOD signal forced during this phase. total soft-start time calculated tSOFTSTART CSS/1.4A When device configured track another supply, feedback voltage other supply duplicated resistor divider applied TRACK/SS pin. Therefore, voltage ramp rate this determined ramp rate other supply output voltage.
Fault Conditions: Current Limit Foldback maximum inductor current inherently limited current mode controller maximum sense voltage. LTC3810, maximum sense voltage controlled voltage VRNG pin. With valley current control, maximum sense voltage sense resistance determine maximum allowed inductor valley current. corresponding output current limit ILIMIT VSNS(MAX) RDS(ON)
current limit value should checked ensure that ILIMIT(MIN) IOUT(MAX). minimum value current limit generally occurs with largest highest ambient temperature, conditions that cause largest power loss converter. Note that important check self-consistency between assumed MOSFET junction temperature resulting value ILIMIT which heats MOSFET switches. Caution should used when setting current limit based upon RDS(ON) MOSFETs. maximum current limit determined minimum MOSFET on-resistance. Data sheets typically specify nominal maximum values RDS(ON), minimum. reasonable assumption that minimum RDS(ON)
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LTC3810 APPLICATIONS INFORMATION
Output Voltage Tracking LTC3810 allows user program output ramps means TRACK/SS pin. Through this pin, output either coincidentally ratiometrically track with another supply's output, shown Figure following discussions, VOUT1 refers master LTC3810's output VOUT2 refers slave LTC3810's output. implement coincident tracking Figure 15a, connect additional resistive divider VOUT1 connect midpoint TRACK/SS slave ratio this divider should selected same that slave IC's feedback divider shown Figure this tracking mode, VOUT1 must higher than VOUT2. implement ratiometric tracking, ratio divider should exactly same master IC's feedback divider. Note that internal soft-start current will introduce small
VOUT1 OUTPUT VOLTAGE OUTPUT VOLTAGE
VOUT1
VOUT2
VOUT2
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TIME
TIME
(15a) Coincident Tracking
(15b) Ratiometric Tracking
Figure Different Modes Output Voltage Tracking
VOUT1 TRACK/SS2 VFB1 VFB2
VOUT2
VOUT1 TRACK/SS2 VFB1 VFB2
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VOUT2
(16a) Coincident Tracking Setup
(16b) Ratiometric Tracking Setup
Figure Setup Coincident Ratiometric Tracking
TRACK/SS2 0.8V VFB2
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Figure Equivalent Input Circuit Error Amplifier
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LTC3810 APPLICATIONS INFORMATION
error tracking voltage depending absolute values tracking resistive divider. selecting different resistors, LTC3810 achieve different modes tracking including Figure which mode should programmed? While either mode Figure satisfies most practical applications, there exist some tradeoffs. ratiometric mode saves pair resistors, coincident mode offers better output regulation. This better understood with help Figure input stage slave IC's error amplifier, common anode diodes used clamp equivalent reference voltage additional diode used match shifted common mode voltage. current sources same amplitude. coincident mode, TRACK/SS voltage substantially higher than 0.8V steady state effectively turns will therefore conduct same current offer tight matching between VFB2 internal precision 0.8V reference. ratiometric mode, however, TRACK/SS equals 0.8V steady state. will divert part bias current make VFB2 slightly lower than 0.8V. Although this error minimized exponential characteristic diode, does impose finite amount output voltage deviation. Furthermore, when master IC's output experiences dynamic excursion (under load transient, example), slave output will affected well. better output regulation, coincident tracking mode instead ratiometric. Phase-Locked Loop Frequency Synchronization LTC3810 phase-locked loop comprised internal voltage controlled oscillator phase detector. This allows MOSFET turn-on locked rising edge external source. frequency range voltage controlled oscillator ±30% around center frequency center frequency operating frequency discussed Operating Frequency section. LTC3810 incorporates pulse detection circuit that will detect clock MODE/SYNC pin. turn, will turn phase-locked loop function. pulse width clock greater than 400ns amplitude clock should greater than
MODE/SYNC DIGITAL PHASE/ FREQUENCY DETECTOR
internal oscillator locks external clock after second clock transition received. When external synchronization detected, LTC3810 will operate forced continuous mode. external clock transition detected three successive periods, internal oscillator will revert frequency programmed resistor. During start-up phase, phase-locked loop function disabled. When LTC3810 synchronization mode, PLL/LPF voltage around 1.215V. Frequency synchronization accomplished changing internal on-time current according voltage PLL/LPF pin. phase detector used edge sensitive digital type which provides zero degrees phase shift between external internal pulses. This type phase detector will lock input frequencies close harmonics center frequency. hold-in range, equal capture range, ±0.3 output phase detector complementary pair current sources charging discharging external filter network PLL/LPF pin. simplified block diagram shown Figure
2.4V PLL/LPF
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Figure Phase-Locked Loop Block Diagram
external frequency (fMODE/SYNC) greater than oscillator frequency current sourced continuously, pulling PLL/LPF pin. When external frequency
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LTC3810 APPLICATIONS INFORMATION
less than current sunk continuously, pulling down PLL/LPF pin. external internal frequencies same exhibit phase difference, current sources turn amount time corresponding phase difference. Thus voltage PLL/LPF adjusted until phase frequency external internal oscillators identical. this stable operating point phase comparator output open filter capacitor holds voltage. LTC3810 MODE/SYNC must driven from impedance source such logic gate located close pin. loop filter components (CLP, RLP) smooth current pulses from phase detector provide stable input voltage controlled oscillator. filter components determine fast loop acquires lock. Typically 0.01F 0.1F Clearance/Creepage Considerations LTC3810 available package which 0.0106" spacing between adjacent pins. maximize board trace clearance between high voltage pins, LTC3810 three unconnected pins between adjacent high voltage voltage pins, providing 4(0.0106") 0.042" clearance which will sufficient most applications 100V. more information, refer printed circuit board design standards described IPC-2221 (www.ipc.org). Efficiency Considerations percent efficiency switching regulator equal output power divided input power times 100%. often useful analyze individual losses determine what limiting efficiency which change would produce most improvement. Although dissipative elements circuit produce losses, four main sources account most losses LTC3810 circuits: losses. These arise from resistances MOSFETs, inductor board traces cause efficiency drop high output currents. continuous mode average output current flows through chopped between bottom MOSFETs. MOSFETs have approximately same RDS(ON), then resistance MOSFET simply summed with resistances board traces obtain loss. example, RDS(ON) 0.01 0.005, loss will range from 15mW 1.5W output current varies from 10A. Transition loss. This loss arises from brief amount time MOSFET spends saturated region during switch node transitions. depends upon input voltage, load current, driver strength MOSFET capacitance, among other factors. loss significant input voltages above estimated from second term PMAIN equation found Power MOSFET Selection section. When transition losses significant, efficiency improved lowering frequency and/or using MOSFET(s) with lower CRSS expense higher RDS(ON). INTVCC/DRVCC current. This MOSFET driver control currents. Control current typically about driver current calculated IGATE f(QG(TOP) QG(BOT)), where QG(TOP) QG(BOT) gate charges bottom MOSFETs. This loss proportional supply voltage that INTVCC/ DRVCC derived from, i.e., external NMOS linear regulator, VOUT internal EXTVCC regulator, VEXT when external supply connected INTVCC/DRVCC. loss. input capacitor difficult filtering large input current regulator. must have very minimize loss sufficient capacitance prevent current from causing additional upstream losses fuses batteries. Other losses, including COUT loss, Schottky diode conduction loss during dead time inductor core loss generally account less than additional loss. When making adjustments improve efficiency, input current best indicator changes efficiency. make change input current decreases, then efficiency increased. there change input current, then there change efficiency.
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LTC3810 APPLICATIONS INFORMATION
Checking Transient Response regulator loop response checked looking load transient response. Switching regulators take several cycles respond step load current. When load step occurs, VOUT immediately shifts amount equal ILOAD (ESR), where effective series resistance COUT. ILOAD also begins charge discharge COUT generating feedback error signal used regulator return VOUT steady-state value. During this recovery time, VOUT monitored overshoot ringing that would indicate stability problem. Design Example design example, take supply with following specifications: (48V nominal), VOUT ±5%, IOUT(MAX) 10A, 250kHz. First, calculate timing resistor with INTVCC: 263k 2.4V 250kHz 76pF guarantee proper current limit worst-case conditions, increase nominal VSNS least 320mV tying VRNG 2V). check current limit acceptable VSNS 320mV, assume junction temperature about 80°C above 70°C ambient (150°C ILIMIT 320mV 11.7A 0.0165
double-check assumed MOSFET: PBOT 11.7A 0.0165 3.8W 70°C 3.8W 20°C/W 146°C Verify that Si7852DP also good choice MOSFET checking power dissipation current limit maximum input voltage, assuming junction temperature 50°C above 70°C ambient (120°C 1.7): PMAIN 11.7A (1.7 0.0165 11.7A 288pF 250kHz 4.7V 4.7V 0.64W 1.75W 2.39W
choose inductor about ripple current maximum VIN: 250kHz With inductor, ripple current will vary from 3.2A (32% 40%) over input supply range. Next, choose bottom MOSFET switch. Since drain MOSFET will full supply voltage 72V(max) plus ringing, choose MOSFET provide margin safety. Si7852DP has: BVDSS RDS(ON) 16.5m(max)/13.5m(nom), 0.007/°C, CMILLER (18.5nC 7nC)/40V 288pF VGS(MILLER) 4.7V, 20°C/W. This yields nominal sense voltage VSNS(NOM) 0.0135 176mV
70°C 2.39W 20°C/W 118°C junction temperature will significantly less nominal current, this analysis shows that careful attention heat sinking board will necessary this circuit. Since VOUT 6.7V, INTVCC/DRVCC voltage generated from VOUT with internal connecting VOUT EXTVCC pin. small SOT23 MOSFET such ZXMN10A07F used pass device fault timeout enabled. Choose RNDRV guarantee that fault timeout enabled when power dissipation exceeds 0.4W (max 70°C ambient). Calculate power dissipation VIN(MIN) 36V: 250kHz 34nC 20mA (36V 10V)(0.02A) 0.52W
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LTC3810 APPLICATIONS INFORMATION
Since power dissipation VIN(MIN) already exceeds 0.4W, calculate RNDRV(MAX) such that fault timeout always enabled: RNDRV 3.5V 83.3k 270A immediate connect components ground plane including SGND PGND LTC3810. several bigger vias power components. compact plane switch node (SW) improve cooling MOSFETs keep down. planes VOUT maintain good voltage filtering keep power losses low. Flood unused areas layers with copper. Flooding with copper will reduce temperature rise power component. connect copper areas (VIN, VOUT, other rail your system). When laying printed circuit board, without ground plane, following checklist ensure proper operation controller. Segregate signal power grounds. smallsignal components should return SGND point which then tied PGND close source Place close controller possible, keeping PGND, traces short. Connect input capacitor(s) close power MOSFETs. This capacitor carries MOSFET current. Keep high dV/dt BOOST nodes away from sensitive small-signal nodes. Connect INTVCC decoupling capacitor CVCC closely INTVCC SGND pins. Connect driver boost capacitor closely BOOST pins. Connect bottom driver decoupling capacitor CDRVCC closely DRVCC BGRTN pins.
choose RNDRV 80.6k. chosen current rating about 85°C. output capacitors chosen 0.018 minimize output voltage changes inductor ripple current load steps. ripple voltage will only: VOUT(RIPPLE) IL(MAX) 0.018 72mV However, load step will cause output change VOUT(STEP) ILOAD 0.018 180mV optional ceramic output capacitor included minimize effect output ripple. complete circuit shown Figure Board Layout Checklist When laying board follow suggested approaches. simple board layout requires dedicated ground plane layer. Also, higher currents, recommended multilayer board help with heat sinking power components. ground plane layer should have traces should close possible layer with power MOSFETs. Place CIN, COUT, MOSFETs, inductor compact area. help have some components bottom side board.
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LTC3810 APPLICATIONS INFORMATION
261k 100pF 80.6k PGOOD 250kHz CLOCK 0.01F RUV1 470k RNDRV 80.6k ZXMN10A07F BAS19 BOOST 0.1F Si7852DP CIN1 100V CIN2 100V PGND
LTC3810
1000pF
PGOOD MODE/SYNC PLL/LPF
SENSE+
SENSE- BGRTN DRVCC INTVCC EXTVCC NDRV CDRVCC 0.1F Si7852DP
VOUT COUT1 270F
SS/TRACK
SHDN
SGND SHDN UVIN RUV2 RFB2 47pF 200k SGND
B1100 CVCC
COUT2
PGND
RFB1
3810
Figure Input Voltage 12V/10A Synchronized 250kHz
TYPICAL APPLICATIONS
Input Voltage 5V/5A with Power from Supply Ceramic Output Capacitors
CIN1 100V CIN2 100V PGND 0.1F Si7852DP
110k 100pF BAS19 BOOST VRNG PGOOD MODE/SYNC PLL/LPF
LTC3810
SENSE+
PGOOD RUV1 470k 1000pF
SENSE- BGRTN
CDRVCC 0.1F
4.7H
VOUT COUT 6.3V
SS/TRACK
SHDN
SGND SHDN UVIN RUV2 61.9k RFB2 1.89k 200pF 100k SGND
DRVCC INTVCC EXTVCC NDRV
Si7852DP B1100
CVCC
PGND
RFB1
3810 TA04
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LTC3810 TYPICAL APPLICATIONS
Input Voltage 3.3V/5A with Fault Timeout, Pulse Skip Disabled
RNDRV 274k ZVN4210G BAS19 BOOST 1000pF VRNG PGOOD MODE/SYNC PLL/LPF 0.1F Si7852DP CIN1 100V CIN2 100V PGND
71.5k 100pF
LTC3810
SENSE+
PGOOD
SENSE BGRTN CDRVCC 0.1F Si7852DP B1100 CVCC COUT1 270F 6.3V
4.7H
VOUT 3.3V
DRVCC
SHDN
SS/TRACK SGND SHDN UVIN 47pF RFB2 1.89k 200k SGND
INTVCC EXTVCC NDRV
COUT2 6.3V
PGND
RFB1
3810 TA05
PACKAGE DESCRIPTION
Package 28-Lead Plastic SSOP (5.3mm)
(Reference 05-08-1640)
9.90 10.50* (.390 .413)
1.25 ±0.12
7.40 8.20 (.291 .323)
0.42 ±0.03 RECOMMENDED SOLDER LAYOUT 5.00 5.60** (.197 .221)
0.65 (.079)
0.09 0.25 (.0035 .010)
0.55 0.95 (.022 .037)
0.65 (.0256)
NOTE: CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS DIMENSIONS (INCHES) DRAWING SCALE *DIMENSIONS INCLUDE MOLD FLASH. MOLD FLASH SHALL EXCEED .152mm (.006") SIDE **DIMENSIONS INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL EXCEED .254mm (.010") SIDE
0.22 0.38 (.009 .015)
0.05 (.002)
SSOP 0204
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Information furnished Linear Technology Corporation believed accurate reliable. However, responsibility assumed use. Linear Technology Corporation makes representation that interconnection circuits described herein will infringe existing patent rights.
LTC3810 TYPICAL APPLICATION
100V Input Voltage 12V/5A with Trickle Charger Start-Up
CIN1 100V 100V CIN2 100V PGND Si7456DP 261k 100pF BAS19 BOOST 0.1F RNDRV 100k
LTC3810
PGOOD RUV1 470k 1000pF
VRNG PGOOD MODE/SYNC PLL/LPF
SENSE
SENSE- BGRTN DRVCC INTV
VOUT COUT1 270F
CDRVCC 0.1F
Si7456DP B1100
SS/TRACK
SHDN
SGND SHDN UVIN SGND RUV2 RFB2 47pF 200k
COUT2
EXTVCC NDRV CVCC1
CVCC2
PGND
RFB1
3810 TA06
RELATED PARTS
PART NUMBER LT®1074HV/LT1076HV LTC1735 LTC1778 LT3010 LT3430/LT3431 LT3433 LTC3703 LT3800 LTC3810-5 LTC3812-5 LT3824 LT3844 LT3845 DESCRIPTION Monolithic 5A/2A Step-Down DC/DC Converters Synchronous Step-Down DC/DC Controller RSENSESynchronous DC/DC Controller 50mA, Linear Regulator Monolithic 200kHz/500kHz Step-Down Regulator Monolithic Step-Up/Step-Down DC/DC Converter 100V Synchronous DC/DC Controller High Voltage Synchronous Regulator Controller RSENSE Current Mode Controller RSENSE Current Mode Controller High Voltage Step-Down Controller High Voltage Current Mode Controller with Programmable Operating Frequency High Voltage Synchronous Regulator Controller with Adjustable Operating Frequency COMMENTS 60V, TO-220 Packages 3.5V 36V, 0.8V VOUT Current Mode, IOUT 36V, Fast Transient Response, Current Mode, IOUT 1.275V VOUT 60V, Protection Diode Required, 8-Lead MSOP 5.5V 60V, Saturation Switch, 16-Pin SSOP 60V, 500mA Switch, Automatic Step-Up/Step-Down, 100V, 9.3V Gate Drive Supply 60V, IOUT 20A, Current Mode, Onboard Bias Regulator, Burst Mode Operation, 16-Lead TSSOP Package 60V, IOUT 20A, Large Gate Drivers 60V, IOUT 20A, Large Gate Drivers 60V, P-Channel MOSFET, IOUT MSOP-10 Package 60V, IOUT Onboard Bias Regulator, Burst Mode Operation, Sync Capability, 16-Lead TSSOP Package 60V, IOUT 20A, Onboard Regulator, Burst Mode Operation, 16-Lead TSSOP Package
Burst Mode registered trademark Linear Technology Corporation. RSENSE trademark Linear Technology Corporation.
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Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, 95035-7417
(408) 432-1900 FAX: (408) 434-0507
0408 PRINTED
www.linear.com
LINEAR TECHNOLOGY CORPORATION 2007
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