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PRACTICAL DESIGN TECHNIQUES SENSOR SIGNAL CONDITIONING
INTRODUCTION BRIDGE CIRCUITS AMPLIFIERS SIGNAL CONDITIONING STRAIN, FORCE, PRESSURE, FLOW MEASUREMENTS HIGH IMPEDANCE SENSORS POSITION MOTION SENSORS TEMPERATURE SENSORS ADCs SIGNAL CONDITIONING SMART SENSORS HARDWARE DESIGN TECHNIQUES INDEX
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PRACTICAL DESIGN TECHNIQUES SENSOR SIGNAL CONDITIONING
ACKNOWLEDGMENTS
Thanks many technical staff members Analog Devices Engineering Marketing provided invaluable inputs during this project. Particular credit individual authors whose names appear beginning their material. Special thanks Freeman, Walter Jung, Bill Chestnut, Grokulsky thoroughly reviewing material content accuracy. Judith Douville compiled index, printing done Donnelley Sons, Inc.
Walt Kester 1999 Copyright 1999 Analog Devices, Inc. Printed United States America rights reserved. This book, parts thereof, must reproduced form without permission copyright owner. Information furnished Analog Devices, Inc., believed accurate reliable. However, responsibility assumed Analog Devices, Inc., use. Analog Devices, Inc., makes representation that interconnections circuits described herein will infringe existing future patent rights, descriptions contained herein imply granting licenses make, use, sell equipment constructed accordance therewith. Specifications subject change without notice.
ISBN-0-916550-20-6
PRACTICAL DESIGN TECHNIQUES SENSOR SIGNAL CONDITIONING
SECTION INTRODUCTION SECTION BRIDGE CIRCUITS
Bridge Configurations Amplifying Linearizing Bridge Outputs Driving Bridges
SECTION AMPLIFIERS SIGNAL CONDITIONING
Precision Characteristics Amplifier Error Budget Analysis Single Supply Amps Instrumentation Amplifiers Chopper Stabilized Amplifiers Isolation Amplifiers
SECTION STRAIN, FORCE, PRESSURE, FLOW MEASUREMENTS
Strain Gages Bridge Signal Conditioning Circuits
SECTION HIGH IMPEDANCE SENSORS
Photodiode Preamplifier Design Compensation High Speed Photodiode Converter High Impedance Charge Output Sensors CCD/CIS Image Processing
SECTION POSITION MOTION SENSORS
Linear Variable Differential Transformers (LVDTs) Hall Effect Magnetic Sensors Optical Encoders Resolvers Synchros Inductosyns Vector Induction Motor Control Accelerometers
SECTION TEMPERATURE SENSORS
Thermocouple Principles Cold-Junction Compensation Resistance Temperature Detectors (RTDs) Thermistors Semiconductor Temperature Sensors Microprocessor Temperature Monitoring
SECTION ADCs SIGNAL CONDITIONING
Successive Approximation ADCs ADCs With Multiplexed Inputs Complete Data Acquisition Systems Chip Sigma-Delta Measurement ADCs High Resolution, Low-Frequency Sigma-Delta Measurement ADCs Applications Sigma-Delta ADCs Power Meters
SECTION SMART SENSORS
4-20mA Control Loops Interfacing Sensors Networks MicroConverter
SECTION HARDWARE DESIGN TECHNIQUES
Resistor Thermocouple Errors High Accuracy Systems Grounding Mixed Signal Systems Power Supply Noise Reduction Filtering Preventing Rectification Dealing With High Speed Logic Review Shielding Concepts Isolation Techniques Overvoltage Protection Electrostatic Discharge (ESD)
INDEX
PRACTICAL DESIGN TECHNIQUES SENSOR SIGNAL CONDITIONING
INTRODUCTION BRIDGE CIRCUITS AMPLIFIERS SIGNAL CONDITIONING STRAIN, FORCE, PRESSURE, FLOW MEASUREMENTS HIGH IMPEDANCE SENSORS POSITION MOTION SENSORS TEMPERATURE SENSORS ADCs SIGNAL CONDITIONING SMART SENSORS HARDWARE DESIGN TECHNIQUES INDEX
INTRODUCTION
SECTION INTRODUCTION Walt Kester
This book deals with sensors associated signal conditioning circuits. topic broad, focus this book concentrate circuit signal processing applications sensors rather than details actual sensors themselves. Strictly speaking, sensor device that receives signal stimulus responds with electrical signal, while transducer converter type energy into another. practice, however, terms often used interchangeably. Sensors their associated circuits used measure various physical properties such temperature, force, pressure, flow, position, light intensity, etc. These properties stimulus sensor, sensor output conditioned processed provide corresponding measurement physical property. will cover possible types sensors, only most popular ones, specifically, those that lend themselves process control data acquisition systems. Sensors operate themselves. They generally part larger system consisting signal conditioners various analog digital signal processing circuits. system could measurement system, data acquisition system, process control system, example. Sensors classified number ways. From signal conditioning viewpoint useful classify sensors either active passive. active sensor requires external source excitation. Resistor-based sensors such thermistors, RTDs (Resistance Temperature Detectors), strain gages examples active sensors, because current must passed through them corresponding voltage measured order determine resistance value. alternative would place devices bridge circuit, however either case, external current voltage required. other hand, passive self-generating) sensors generate their electrical output signal without requiring external voltages currents. Examples passive sensors thermocouples photodiodes which generate thermoelectric voltages photocurrents, respectively, which independent external circuits. should noted that these definitions (active passive) refer need lack thereof) external active circuitry produce electrical output signal from sensor. would seem equally logical consider thermocouple active sense that produces output voltage with external circuitry, however convention industry classify sensor with respect external circuit requirement defined above.
INTRODUCTION
SENSOR OVERVIEW
Sensors: Convert Signal Stimulus (Representing Physical Property) into Electrical Output Transducers: Convert Type Energy into Another Terms often Interchanged Active Sensors Require External Source Excitation: RTDs, Strain-Gages Passive (Self-Generating) Sensors not: Thermocouples, Photodiodes Figure
TYPICAL SENSORS THEIR OUTPUTS
PROPERTY SENSOR ACTIVE/ PASSIVE Temperature Thermocouple Passive Silicon Thermistor Force Pressure Acceleration Position Strain Gage Piezoelectric Active Active Active Active Passive Voltage Voltage/Current Resistance Resistance Resistance Voltage Capacitance Voltage Current OUTPUT
Accelerometer Active LVDT Active Passive
Light Intensity Photodiode
Figure
INTRODUCTION
logical classify sensors with respect physical property sensor designed measure. Thus have temperature sensors, force sensors, pressure sensors, motion sensors, etc. However, sensors which measure different properties have same type electrical output. instance, Resistance Temperature Detector (RTD) variable resistance, resistive strain gauge. Both RTDs strain gages often placed bridge circuits, conditioning circuits therefore quite similar. fact, bridges their conditioning circuits deserve detailed discussion. full-scale outputs most sensors (passive active) relatively small voltages, currents, resistance changes, therefore their outputs must properly conditioned before further analog digital processing occur. Because this, entire class circuits have evolved, generally referred signal conditioning circuits. Amplification, level translation, galvanic isolation, impedance transformation, linearization, filtering fundamental signal conditioning functions which required. Whatever form conditioning takes, however, circuitry performance will governed electrical character sensor output. Accurate characterization sensor terms parameters appropriate application, e.g., sensitivity, voltage current levels, linearity, impedances, gain, offset, drift, time constants, maximum electrical ratings, stray impedances other important considerations spell difference between substandard successful application device, especially cases where high resolution precision, low-level measurements involved. Higher levels integration allow play significant role both analog digital signal conditioning. ADCs specifically designed measurement applications often contain on-chip programmable-gain amplifiers (PGAs) other useful circuits, such current sources driving RTDs, thereby minimizing external conditioning circuit requirements. Most sensor outputs non-linear with respect stimulus, their outputs must linearized order yield correct measurements. Analog techniques used perform this function, however recent introduction high performance ADCs allows linearization done much more efficiently accurately software eliminates need tedious manual calibration using multiple sometimes interactive trimpots. application sensors typical process control system shown Figure 1.3. Assume physical property controlled temperature. output temperature sensor conditioned then digitized ADC. microcontroller host computer determines temperature above below desired value, outputs digital word digital-to-analog converter (DAC). output conditioned drives actuator, this case heater. Notice that interface between control center remote process industry-standard 4-20mA loop.
INTRODUCTION
TYPICAL INDUSTRIAL PROCESS CONTROL LOOP
REMOTE
SIGNAL CONDITIONING 20mA TRANSMITTER
CONTROL ROOM
20mA RECEIVER SIGNAL CONDITIONING
TEMP SENSOR
PROCESS
HOST COMPUTER
MICRO CONTROLLER
HEATER
SIGNAL CONDITIONING
20mA RECEIVER
20mA TRANSMITTER
SIGNAL CONDITIONING
Figure
Digital techniques becoming more more popular processing sensor outputs data acquisition, process control, measurement. 8-bit microcontrollers (8051-based, example) generally have sufficient speed processing capability most applications. including conversion microcontroller programmability sensor itself, "smart sensor" implemented with self contained calibration linearization features among others. smart sensor then interface directly industrial network shown Figure 1.4.
basic building blocks "smart sensor" shown Figure 1.5, constructed with multiple ICs. Analog Devices MicroConverter-series products includes on-chip high performance multiplexers, ADCs, DACs, coupled with FLASH Memory industry-standard 8052 microcontroller core, well support circuitry several standard serial port configurations. These first integrated circuits which truly smart sensor data acquisition systems (highperformance data conversion circuits, microcontroller, FLASH memory) single chip (see Figure 1.6).
INTRODUCTION
STANDARDIZATION DIGITAL INTERFACE USING SMART SENSORS
BRANCH
FIELD NETWORK
NODE SMART SENSOR
NODE SMART SENSOR
DEVICE NETWORK
NODE
SMART SENSOR
SMART SENSOR
SMART SENSORS OFFER: Self-Calibration Linearization Interchangeability Standard Digital Interfaces
NODE
Figure
BASIC ELEMENTS "SMART" SENSOR
Pressure Sensor, RTD, Thermocouple, Strain Gage, etc.
Precision Amplifier High Resolution Microcontroller
Figure
Sensor
INTRODUCTION
EVEN SMARTER SENSOR
Pressure Sensor, RTD, Thermocouple, Strain Gage, etc.
MicroConverter
Figure
Sensor
BRIDGE CIRCUITS
SECTION BRIDGE CIRCUITS
Walt Kester INTRODUCTION
This section discusses fundamental concepts bridge circuits, followed section precision amps (Section Section focuses detailed application circuits relating strain gage-based sensors. Sections read sequentially reader already understands design issues relating amps which covered Section Resistive elements some most common sensors. They inexpensive manufacture relatively easy interface with signal conditioning circuits. Resistive elements made sensitive temperature, strain pressure flex), light. Using these basic elements, many complex physical phenomena measured; such fluid mass flow sensing temperature difference between calibrated resistances) dew-point humidity measuring different temperature points), etc. Sensor elements' resistances range from less than several hundred depending sensor design physical environment measured (See Figure 2.1). example, RTDs (Resistance Temperature Devices) typically 1000. Thermistors typically 3500 higher.
RESISTANCE POPULAR SENSORS
Strain Gages Weigh-Scale Load Cells Pressure Sensors Relative Humidity Resistance Temperature Devices (RTDs) Thermistors 120, 350, 3500 3500 3500 100k 1000
Figure
BRIDGE CIRCUITS Resistive sensors such RTDs strain gages produce small percentage changes resistance response change physical variable such temperature force. Platinum RTDs have temperature coefficient about 0.385%/°C. Thus, order accurately resolve temperature measurement accuracy must much better than 0.385 RTD. Strain gages present significant measurement challenge because typical change resistance over entire operating range strain gage less than nominal resistance value. Accurately measuring small resistance changes therefore critical when applying resistive sensors. technique measuring resistance (shown Figure 2.2) force constant current through resistive sensor measure voltage output. This requires both accurate current source accurate means measuring voltage. change current will interpreted resistance change. addition, power dissipation resistive sensor must small, accordance with manufacturer's recommendations, that self-heating does produce errors, therefore drive current must small.
MEASURING RESISTANCE INDIRECTLY USING CONSTANT CURRENT SOURCE
VOUT
Figure
Bridges offer attractive alternative measuring small resistance changes accurately. basic Wheatstone bridge (actually developed Christie 1833) shown Figure 2.3. consists four resistors connected form quadrilateral, source excitation (voltage current) connected across diagonals, voltage detector connected across other diagonal. detector measures difference between outputs voltage dividers connected across excitation.
BRIDGE CIRCUITS
WHEATSTONE BRIDGE
BALANCE,
Figure bridge measures resistance indirectly comparison with similar resistance. principle ways operating bridge null detector device that reads difference directly voltage. When R1/R4 R2/R3, resistance bridge null, irrespective mode excitation (current voltage, DC), magnitude excitation, mode readout (current voltage), impedance detector. Therefore, ratio R2/R3 fixed null achieved when unknown accurately determined variable resistance, magnitude found adjusting until null achieved. Conversely, sensor-type measurements, fixed reference, null occurs when magnitude external variable (strain, temperature, etc.) such that Null measurements principally used feedback systems involving electromechanical and/or human elements. Such systems seek force active element (strain gage, RTD, thermistor, etc.) balance bridge influencing parameter being measured. majority sensor applications employing bridges, however, deviation more resistors bridge from initial value measured indication magnitude change) measured variable. this case, output voltage change indication resistance change. Because very small resistance changes common, output voltage change small tens millivolts, even with typical excitation voltage load cell application).
BRIDGE CIRCUITS many bridge applications, there two, even four elements which vary. Figure shows four commonly used bridges suitable sensor applications corresponding equations which relate bridge output voltage excitation voltage bridge resistance values. this case, assume constant voltage drive, Note that since bridge output directly proportional measurement accuracy better than that accuracy excitation voltage.
OUTPUT VOLTAGE LINEARITY ERROR CONSTANT VOLTAGE DRIVE BRIDGE CONFIGURATIONS
0.5%/%
0.5%/%
Linearity Error:
Single-Element Varying
Two-Element Varying Figure
Two-Element All-Element Varying Varying
each case, value fixed bridge resistor, chosen equal nominal value variable resistor(s). deviation variable resistor(s) about nominal value proportional quantity being measured, such strain case strain gage) temperature case RTD). sensitivity bridge ratio maximum expected change output voltage excitation voltage. instance, 10V, fullscale bridge output 10mV, then sensitivity 1mV/V. single-element varying bridge most suited temperature sensing using RTDs thermistors. This configuration also used with single resistive strain gage. resistances nominally equal, them (the sensor) variable amount equation indicates, relationship between bridge output linear. example, (0.1%
BRIDGE CIRCUITS change resistance), output bridge 2.49875mV 10V. error 2.50000mV 2.49875mV, 0.00125mV. Converting this fullscale dividing 2.5mV yields end-point linearity error percent approximately 0.05%. (Bridge end-point linearity error calculated worst error from straight line which connects origin point i.e. gain error included). change resistance), output bridge 24.8756mV, representing end-point linearity error approximately 0.5%. end-point linearity error single-element bridge expressed equation form: Single-Element Varying Bridge End-Point Linearity Error Change Resistance should noted that above nonlinearity refers nonlinearity bridge itself sensor. practice, most sensors exhibit certain amount their nonlinearity which must accounted final measurement. some applications, bridge nonlinearity acceptable, there various methods available linearize bridges. Since there fixed relationship between bridge resistance change output (shown equations), software used remove linearity error digital systems. Circuit techniques also used linearize bridge output directly, these will discussed shortly. There possibilities consider case two-element varying bridge. first, Case (1), both elements change same direction, such identical strain gages mounted adjacent each other with their axes parallel. nonlinearity same that single-element varying bridge, however gain twice that single-element varying bridge. two-element varying bridge commonly found pressure sensors flow meter systems. second configuration two-element varying bridge, Case (2), requires identical elements that vary opposite directions. This could correspond identical strain gages: mounted flexing surface, bottom. Note that this configuration linear, like two-element Case (1), twice gain single-element configuration. Another view this configuration consider terms comprising sections center-tapped potentiometer. all-element varying bridge produces most signal given resistance change inherently linear. industry-standard configuration load cells which constructed from four identical strain gages. Bridges also driven from constant current sources shown Figure 2.5. Current drive, although popular voltage drive, advantage when bridge located remotely from source excitation because wiring resistance does introduce errors measurement. Note also that with constant current excitation, configurations linear with exception single-element varying case.
BRIDGE CIRCUITS
OUTPUT VOLTAGE LINEARITY ERROR CONSTANT CURRENT DRIVE BRIDGE CONFIGURATIONS
Linearity Error:
0.25%/%
Single-Element Varying
Two-Element Varying
Two-Element All-Element Varying Varying
Figure
summary, there many design issues relating bridge circuits. After selecting basic configuration, excitation method must determined. value excitation voltage current must first determined. Recall that fullscale bridge output directly proportional excitation voltage current). Typical bridge sensitivites 1mV/V 10mV/V. Although large excitation voltages yield proportionally larger fullscale output voltages, they also result higher power dissipation possibility sensor resistor self-heating errors. other hand, values excitation voltage require more gain conditioning circuits increase sensitivity noise.
Regardless value, stability excitation voltage current directly affects overall accuracy bridge output. Stable references and/or ratiometric techniques required maintain desired accuracy.
BRIDGE CIRCUITS
BRIDGE CONSIDERATIONS
Selecting Configuration Element Varying) Selection Voltage Current Excitation Stability Excitation Voltage Current Bridge Sensitivity: Output Excitation Voltage 10mV Typical Fullscale Bridge Outputs: 10mV 100mV Typical Precision, Noise Amplification Conditioning Techniques Required Linearization Techniques Required Remote Sensors Present Challenges
Figure
AMPLIFYING LINEARIZING BRIDGE OUTPUTS
output single-element varying bridge amplified single precision op-amp connected inverting mode shown Figure 2.7. This circuit, although simple, poor gain accuracy also unbalances bridge loading from bias current. resistors must carefully chosen matched maximize common mode rejection (CMR). Also difficult maximize while same time allowing different gain options. addition, output nonlinear. redeeming feature circuit that capable single supply operation requires single amp. Note that resistor connected non-inverting input returned VS/2 (rather than ground) that both positive negative values accommodated, output referenced VS/2. much better approach instrumentation amplifier (in-amp) shown Figure 2.8. This efficient circuit provides better gain accuracy (usually with single resistor, does unbalance bridge. Excellent common mode rejection achieved with modern in-amps. bridge's intrinsic characteristics, output nonlinear, this corrected software (assuming that in-amp output digitized using analog-to-digital converter followed microcontroller microprocessor). Instrumentation amplifiers such AD620, AD623, AD627 used single supply applications provided restrictions gain input output voltage swings observed. (For detailed discussion these important considerations, Section
BRIDGE CIRCUITS
USING SINGLE BRIDGE AMPLIFIER SINGLE-ELEMENT VARYING BRIDGE
Figure
USING INSTRUMENTATION AMPLIFIER WITH SINGLE-ELEMENT VARYING BRIDGE
VOUT
GAIN
VOUT
-VS*
TEXT REGARDING
SINGLE-SUPPLY OPERATION
Figure
BRIDGE CIRCUITS Various techniques available linearize bridges, important distinguish between linearity bridge equation linearity sensor response phenomenon being sensed. example, active element RTD, bridge used implement measurement might have perfectly adequate linearity; output could still nonlinear RTD's nonlinearity. Manufacturers sensors employing bridges address nonlinearity issue variety ways, including keeping resistive swings bridge small, shaping complimentary nonlinear response into active elements bridge, using resistive trims first-order corrections, others. Figure shows single-element varying active bridge which produces forced null, adding voltage series with variable arm. That voltage equal magnitude opposite polarity incremental voltage across varying element linear with Since output, used impedance output point bridge measurement. This active bridge gain over standard single-element varying bridge, output linear, even large values Because small output signal, this bridge must usually followed second amplifier. amplifier used this circuit requires dual supplies because output must negative.
LINEARIZING SINGLE-ELEMENT VARYING BRIDGE METHOD
Another circuit linearizing single-element varying bridge shown Figure 2.10. bottom bridge driven amp, which maintains constant current varying resistance element. output signal taken from righthand bridge amplified non-inverting amp. output linear, circuit requires amps which must operate dual supplies. addition, must matched accurate gain.
VOUT
Figure
BRIDGE CIRCUITS
LINEARIZING SINGLE-ELEMENT VARYING BRIDGE METHOD
VOUT
VOUT
VOUT
Figure 2.10 circuit linearizing voltage-driven two-element varying bridge shown Figure 2.11. This circuit similar Figure twice sensitivity. dual supply required. Additional gain necessary.
LINEARIZING TWO-ELEMENT VARYING BRIDGE METHOD (CONSTANT VOLTAGE DRIVE)
Figure 2.11 2.10
BRIDGE CIRCUITS
two-element varying bridge circuit Figure 2.12 uses amp, sense resistor, voltage reference maintain constant current through bridge VREF/RSENSE). current through each bridge remains constant (IB/2) resistances change, therefore output linear function instrumentation amplifier provides additional gain. This circuit operated single supply with proper choice amplifiers signal levels.
LINEARIZING TWO-ELEMENT VARYING BRIDGE METHOD (CONSTANT CURRENT DRIVE)
RSENSE -VS* VREF -VS* VOUT VOUT GAIN
TEXT REGARDING SINGLE-SUPPLY OPERATION
Figure 2.12
DRIVING BRIDGES
Wiring resistance noise pickup biggest problems associated with remotely located bridges. Figure 2.13 shows strain gage which connected rest bridge circuit feet gage twisted pair copper wire. resistance wire 0.105/ft, 10.5 100ft. total lead resistance series with strain gage therefore temperature coefficient copper wire will calculate gain offset error bridge output temperature rise cable. These calculations easy make, because bridge output voltage simply difference between output voltage dividers, each driven from +10V source.
2.11
BRIDGE CIRCUITS
ERRORS PRODUCED WIRING RESISTANCE REMOTE RESISTIVE BRIDGE SENSOR
+10V FEET, GAGE COPPER WIRE 10.5 25°C 0.385%/°C ASSUME +10°C TEMPERATURE CHANGE NUMBERS +35°C
RLEAD 10.5 (10.904) STRAIN GAGE 353.5 RLEAD 10.5 (10.904)
23.45mV (5.44mV 28.83mV)
RCOMP
OFFSET ERROR OVER TEMPERATURE +23%FS GAIN ERROR OVER TEMPERATURE -0.26%FS
Figure 2.13 fullscale variation strain gage resistance (with flex) above nominal value (+3.5), corresponding fullscale strain gage resistance 353.5 which causes bridge output voltage +23.45mV. Notice that addition RCOMP resistor compensates wiring resistance balances bridge when strain gage resistance 350. Without RCOMP, bridge would have output offset voltage 145.63mV nominal strain gage resistance 350. This offset could compensated software just easily, this example, chose with RCOMP. Assume that cable temperature increases above nominal room temperature. This results total lead resistance increase +0.404 each lead. Note: values parentheses diagram indicate values total additional lead resistance leads) +0.808. With strain, this additional lead resistance produces offset +5.44mV bridge output. Fullscale strain produces bridge output +28.83mV change +23.39mV from strain). Thus increase temperature produces offset voltage error +5.44mV (+23% fullscale) gain error -0.06mV (23.39mV 23.45mV), -0.26% fullscale. Note that these errors produced solely gage wire, include temperature coefficient errors strain gage itself. effects wiring resistance bridge output minimized 3-wire connection shown Figure 2.14. assume that bridge output voltage measured high impedance device, therefore there current sense lead. Note that sense lead measures voltage output divider: half bridge resistor plus lead resistance, bottom half strain gage resistance plus lead resistance. nominal sense voltage therefore 2.12
BRIDGE CIRCUITS independent lead resistance. When strain gage resistance increases fullscale (353.5), bridge output increases +24.15mV. Increasing temperature increases lead resistance +0.404 each half divider. fullscale bridge output voltage decreases +24.13mV because small loss sensitivity, there offset error. gain error temperature increase therefore only -0.02mV, -0.08% fullscale. Compare this +23% fullscale offset error -0.26% gain error two-wire connection shown Figure 2.13.
3-WIRE CONNECTION REMOTE BRIDGE ELEMENT (SINGLE-ELEMENT VARYING)
+10V FEET, GAGE COPPER WIRE 10.5 25°C 0.385%/°C ASSUME +10°C TEMPERATURE CHANGE NUMBERS +35°C
RLEAD 10.5 (10.904) STRAIN GAGE 353.5 RLEAD 10.5 (10.904)
24.15mV 24.13mV)
OFFSET ERROR OVER TEMPERATURE 0%FS GAIN ERROR OVER TEMPERATURE -0.08%FS
Figure 2.14 three-wire method works well remotely located resistive elements which make single-element varying bridge. However, all-element varying bridges generally housed complete assembly, case load cell. When these bridges remotely located from conditioning electronics, special techniques must used maintain accuracy. particular concern maintaining accuracy stability bridge excitation voltage. bridge output directly proportional excitation voltage, drift excitation voltage produces corresponding drift output voltage. this reason, most all-element varying bridges (such load cells) six-lead assemblies: leads bridge output, leads bridge excitation, sense leads. This method (called Kelvin 4-wire sensing) shown Figure 2.15. sense lines high impedance inputs, thus there minimal error bias current induced voltage drop across their lead resistance. amps maintain required excitation voltage make voltage measured between sense leads always equal Although Kelvin sensing eliminates 2.13
BRIDGE CIRCUITS errors voltage drops wiring resistance, drive voltages must still highly stable since they directly affect bridge output voltage. addition, amps must have offset, drift, noise.
KELVIN (4-WIRE) SENSING MINIMIZES ERRORS LEAD RESISTANCE
+FORCE RLEAD 6-LEAD BRIDGE +SENSE
SENSE RLEAD
FORCE
Figure 2.15 constant current excitation method shown Figure 2.16 another method minimizing effects wiring resistance measurement accuracy. However, accuracy reference, sense resistor, influence overall accuracy. very powerful ratiometric technique which includes Kelvin sensing minimize errors wiring resistance also eliminates need accurate excitation voltage shown Figure 2.17. AD7730 measurement driven from single supply voltage which also used excite remote bridge. Both analog input reference input high impedance fully differential. using SENSE outputs from bridge differential reference ADC, there loss measurement accuracy actual bridge excitation voltage varies. AD7730 family sigma-delta ADCs with high resolution bits) internal programmable gain amplifiers (PGAs) ideally suited bridge applications. These ADCs have self- system calibration features which allow offset gain errors minimized. instance, AD7730 offset drift gain drift Offset gain errors reduced microvolts using system calibration feature. more detailed discussion these ADCs found Section
2.14
BRIDGE CIRCUITS
CONSTANT CURRENT EXCITATION MINIMIZES WIRING RESISTANCE ERRORS
RLEAD 4-LEAD BRIDGE VREF
RLEAD VREF RSENSE RSENSE
Figure 2.16
DRIVING REMOTE BRIDGE USING KELVIN (4-WIRE) SENSING RATIOMETRIC CONNECTION
+FORCE RLEAD 6-LEAD BRIDGE +SENSE AVDD VREF VREF +5V/+3V DVDD
AD7730 BITS
SENSE
FORCE
RLEAD
Figure 2.17
2.15
BRIDGE CIRCUITS
Maintaining accuracy 0.1% better with fullscale bridge output voltage 20mV requires that offset errors less than 20µV. Figure 2.18 shows some typical sources offset error that inevitable system. Parasitic thermocouples whose junctions different temperatures generate voltages between tens microvolts temperature differential. diagram shows typical parasitic junction formed between copper printed circuit board traces kovar pins amplifier. This thermocouple voltage about temperature differential. thermocouple voltage significantly less when using plastic package with copper lead frame.
amplifier offset voltage bias current other sources offset error. amplifier bias current must flow through source impedance. unbalance either source resistances bias currents produce offset errors. addition, offset voltage bias currents function temperature. High performance offset, offset drift, bias current, noise precision amplifiers such OP177 AD707 required. some cases, chopper-stabilized amplifiers such AD8551/AD8552/AD8554 only solution.
TYPICAL SOURCES OFFSET VOLTAGE
THERMOCOUPLE VOLTAGE 35µV/ COPPER TRACES KOVAR PINS
Figure 2.18
2.16
BRIDGE CIRCUITS
bridge excitation shown Figure 2.19 effectively remove offset voltages series with bridge output. concept simple. bridge output voltage measured under conditions shown. first measurement yields measurement where desired bridge output voltage offset error voltage EOS. polarity bridge excitation reversed, second measurement made. Subtracting from yields 2VO, offset error term cancels shown. Obviously, this technique requires highly accurate measurement (such AD7730) well microcontroller perform subtraction. ratiometric reference desired, must also accommodate changing polarity reference voltage. Again, AD7730 includes this capability.
EXCITATION MINIMIZES OFFSET ERRORS
NORMAL DRIVE VOLTAGES OFFSET ERRORS
EOS) EOS) REVERSE DRIVE VOLTAGES
Figure 2.19
P-Channel N-Channel MOSFETs configured bridge driver shown Figure 2.20. Dedicated bridge driver chips also available, such Micrel MIC4427. Note that because on-resistance MOSFETs, Kelvin sensing must used these applications. also important that drive signals non-overlapping prevent excessive MOSFET switching currents. AD7730 chip circuitry generate required non-overlapping drive signals excitation.
2.17
BRIDGE CIRCUITS
SIMPLIFIED BRIDGE DRIVE CIRCUIT
V3,4
SENSE
SENSE
V1,2
V1,2 V3,4
Q1,Q2 Q3,Q4
Q1,Q2 Q3,Q4
Figure 2.20
2.18
BRIDGE CIRCUITS
REFERENCES
Ramon Pallas-Areny John Webster, Sensors Signal Conditioning, John Wiley, York, 1991. Sheingold, Editor, Transducer Interfacing Handbook, Analog Devices, Inc., 1980. Walt Kester, Editor, 1992 Amplifier Applications Guide, Section Analog Devices, Inc., 1992. Walt Kester, Editor, System Applications Guide, Section Analog Devices, Inc., 1993. AD7730 Data Sheet, Analog Devices, available http://www.analog.com.
2.19
AMPLIFIERS SIGNAL CONDITIONING
SECTION AMPLIFIERS SIGNAL CONDITIONING
Walt Kester, James Bryant, Walt Jung INTRODUCTION
This section examines critical parameters amplifiers precision signal conditioning applications. Offset voltages precision amps 10µV with corresponding temperature drifts Chopper stabilized amps provide offsets offset voltage drifts which cannot distinguished from noise. Open loop gains greater than million common, along with common mode power supply rejection ratios same magnitude. Applying these precision amplifiers while maintaining amplifier performance present significant challenges design engineer, i.e., external passive component selection board layout. important understand that open-loop gain, offset voltage, power supply rejection (PSR), common mode rejection (CMR) alone should only considerations selecting precision amplifiers. performance amplifier also important, even "low" frequencies. Open-loop gain, PSR, have relatively corner frequencies, therefore what considered "low" frequency actually fall above these corner frequencies, increasing errors above value predicted solely parameters. example, amplifier having open-loop gain million unity-gain crossover frequency 1MHz corresponding corner frequency 0.1Hz! must therefore consider open loop gain actual signal frequency. relationship between single-pole unitygain crossover frequency, signal frequency, fsig, open-loop gain AVOL(fsig) (measured signal frequency given VOL( fsig fsig
example above, open loop gain 100kHz, 100,000 10Hz. Loss open loop gain frequency interest introduce distortion, especially audio frequencies. Loss line frequency harmonics thereof also introduce errors. challenge selecting right amplifier particular signal conditioning application been complicated sheer proliferation various types amplifiers various processes (Bipolar, Complementary Bipolar, BiFET, CMOS, BiCMOS, etc.) architectures (traditional amps, instrumentation amplifiers, chopper amplifiers, isolation amplifiers, etc.) addition, wide selection precision amplifiers available which operate single supply voltages which complicates design process even further because reduced signal swings voltage input output restrictions. Offset voltage noise more significant portion input signal. Selection guides parametric search engines which simplify this process somewhat available world-wideweb (http://www.analog.com) well CDROM.
AMPLIFIERS SIGNAL CONDITIONING
this section, will first look some performance specifications precision amps. Other amplifiers will then examined such instrumentation amplifiers, chopper amplifiers, isolation amplifiers. implications single supply operation will discussed detail because their significance today's designs, which often operate from batteries other power sources.
AMPLIFIERS SIGNAL CONDITIONING
Input Offset Voltage Input Offset Voltage Drift Input Bias Current Input Offset Current Open Loop Gain Unity Gain Bandwidth Product, (0.1Hz 10Hz) Noise Wideband Noise CMR, Single Supply Operation Power Dissipation Figure <100µV <1µV/°C <2nA <2nA >1,000,000 500kHz 5MHz
Always Check Open Loop Gain Signal Frequency! <1µV <10nV/Hz >100dB
PRECISION CHARACTERISTICS
Input Offset Voltage
Input offset voltage error usually largest error sources precision amplifier circuit designs. However, systemic error usually dealt with using manual offset null trim system calibration techniques using microcontroller microprocessor. Both solutions carry cost penalty, today's precision amps offer initial offset voltages 10µV bipolar devices, less chopper stabilized amplifiers. With offset amplifiers, possible eliminate need manual trims system calibration routines.
AMPLIFIERS SIGNAL CONDITIONING Measuring input offset voltages microvolts requires that test circuit does introduce more error than offset voltage itself. Figure shows circuit measuring offset voltage. circuit amplifies input offset voltage noise gain (1001). measurement made amplifier output using accurate digital voltmeter. offset referred input (RTI) calculated dividing output voltage noise gain. small source resistance seen R1||R2 results negligible bias current contribution measured offset voltage. example, bias current flowing through resistor produces 0.02µV error referred input.
MEASURING INPUT OFFSET VOLTAGE
VOUT
VOUT VOUT 1001
OP177A: 10µV maximum DRIFT 0.1µV/°C maximum STABILITY 0.2µV/month typical Figure
simple looks, this circuit give inaccurate results. largest potential source error comes from parasitic thermocouple junctions formed where different metals joined. thermocouple voltage formed temperature difference between junctions range from more than Note that circuit additional resistors have been added non-inverting input order exactly match thermocouple junctions inverting input path. accuracy measurement depends mechanical layout components they placed board. Keep mind that connections component such resistor create equal, opposite polarity thermoelectric voltages (assuming they connected same metal, such copper trace board) which cancel each other assuming both exactly
AMPLIFIERS SIGNAL CONDITIONING same temperature. Clean connections short lead lengths help minimize temperature gradients increase accuracy measurement. Airflow should minimal that thermocouple junctions stabilize same temperature. some cases, circuit should placed small closed container eliminate effects external currents. circuit should placed flat surface that convection currents flow board, across components would case board mounted vertically. Measuring offset voltage shift over temperature even more demanding challenge. Placing printed circuit board containing amplifier being tested small plastic with foam insulation prevents temperature chamber current from causing thermal gradients across parasitic thermocouples. cold testing required, nitrogen purge recommended. Localized temperature cycling amplifier itself using Thermostream-type heater/cooler alternative, however these units tend generate quite airflow which troublesome. addition temperature related drift, offset voltage amplifier changes time passes. This aging effect generally specified long-term stability µV/month, µV/1000 hours, this misleading. Since aging "drunkard's walk" phenomenon, proportional square root elapsed time. aging rate 1µV/1000 hours becomes about 3µV/year, 9µV/year. Long-term stability OP177 AD707 approximately 0.3µV/month. This refers time period after first days operation. Excluding initial hour operation, changes offset voltage these devices during first days operation typically less than 2µV. general rule thumb, prudent control amplifier offset voltage device selection whenever possible, sometimes trim desired. Many precision amps have pins available optional offset null. Generally, pins joined potentiometer, wiper goes supplies through resistor shown Figure 3.3. wiper connected wrong supply, will probably destroyed, data sheet instructions must carefully observed! range offset adjustment precision should more than three times maximum offset voltage lowest grade device, order minimize sensitivity these pins. voltage gain between offset adjustment pins output actually greater than gain signal inputs! therefore very important keep these pins free noise. inadvisable have long leads from remote potentiometer. minimize offset error supply current, connect directly pertinent device supply pin, such shown diagram.
AMPLIFIERS SIGNAL CONDITIONING
OP177/AD707 OFFSET ADJUSTMENT PINS
10k, OFFSET ADJUST RANGE 200µV OFFSET ADJUST RANGE
Figure important note that offset drift with temperature will vary with setting offset adjustment. most cases bipolar will have minimum drift minimum offset. offset adjustment pins should therefore used only adjust amp's offset, correct system offset errors, since this would expense increased temperature drift. drift penalty JFET input much worse than bipolar input order each millivolt nulled offset voltage. generally better control offset voltage proper selection devices device grades. Dual, triple, quad, single amps small packages generally have null capability because count limitations, offset adjustments must done elsewhere system when using these devices. This accomplished with minimal impact drift universal trim, which sums small voltage into input.
Input Offset Voltage Input Bias Current Models
Thus far, have considered only input offset voltage. However, input bias currents also contribute offset error shown generalized model Figure 3.4. useful refer offsets input (RTI) that they easily compared with input signal. equations diagram given total offset voltage referred input (RTI) referred output (RTO).
AMPLIFIERS SIGNAL CONDITIONING
TOTAL OFFSET VOLTAGE MODEL
GAIN FROM OUTPUT
NOISE GAIN
VOUT
GAIN FROM OUTPUT
OFFSET (RTO) OFFSET (RTI
BIAS CURRENT CANCELLATION: OFFSET (RTI)
Figure
precision having standard bipolar input stage using either PNPs NPNs, input bias currents typically 50nA 400nA well matched. making equal parallel combination their effect offset voltage approximately canceled, thus leaving offset current, i.e., difference between input currents error. This current usually order magnitude lower than bias current specification. This scheme, however, does work bias-current compensated bipolar amps (such OP177 AD707) shown Figure 3.5. Bias-current compensated input stages have most good features simple bipolar input stage: offset drift, voltage noise. Their bias current fairly stable over temperature. additional current sources reduce bias currents typically between 0.5nA 10nA. However, signs input bias currents same, they well matched, very low. Typically, specification offset current (the difference between input bias currents) bias-current compensated amps generally about same individual bias currents. case standard bipolar differential pair with bias-current compensation, offset current specification typically five times lower than bias current specification.
AMPLIFIERS SIGNAL CONDITIONING
INPUT BIAS CURRENT COMPENSATED AMPS
UNCOMPENSATED COMPENSATED
MATCHED BIAS CURRENTS SAME SIGN 50nA 10µA 50pA (Super Beta) IOFFSET IBIAS
LOW, UNMATCHED BIAS CURRENTS HAVE DIFFERENT SIGNS 0.5nA 10nA HIGHER CURRENT NOISE IOFFSET IBIAS
Figure
Open Loop Gain Nonlinearity
well understood that order maintain accuracy, precision amplifier's open loop gain, AVOL, should high. This seen examining equation closed loop gain: Noise gain (NG) simply gain seen small voltage source series with input also amplifier signal gain noninverting mode. AVOL above equation infinite, closed loop gain exactly equal noise gain. However, finite values AVOL, there closed loop gain error given equation: Closed Loop Gain %Gain Error 100% 100% AVOL.
Notice from equation that percent gain error directly proportional noise gain, therefore effects finite AVOL less gain. first example Figure where noise gain 1000 shows that open loop gain million, there gain error about 0.05%. open loop gain stays constant over temperature various output loads voltages, gain error calibrated measurement, there then overall system gain error. however, open loop gain changes, closed loop gain will also change, thereby introducing gain uncertainty. second example figure, AVOL decrease 300,000 produces gain error 0.33%, introducing gain
AMPLIFIERS SIGNAL CONDITIONING uncertainty 0.28% closed loop gain. most applications, using proper amplifier, resistors around circuit will largest source gain error.
CHANGES OPEN LOOP GAIN CAUSE CLOSED LOOP GAIN UNCERTAINTY
"IDEAL" CLOSED LOOP GAIN NOISE GAIN ACTUAL CLOSED LOOP GAIN CLOSED LOOP GAIN ERROR
100% 100%
Assume AVOL 2,000,000, 1,000 %GAIN ERROR 0.05% Assume AVOL Drops 300,000 %GAIN ERROR 0.33% CLOSED LOOP GAIN UNCERTAINTY 0.33% 0.05% 0.28% Figure Changes output voltage level output loading most common causes changes open loop gain amps. change open loop gain with signal level produces nonlinearity closed loop gain transfer function which cannot removed during system calibration. Most amps have fixed loads, AVOL changes with load generally important. However, sensitivity AVOL output signal level increase higher load currents. severity nonlinearity varies widely from device type device type, generally specified data sheet. minimum AVOL always specified, choosing with high AVOL will minimize probability gain nonlinearity errors. Gain nonlinearity come from many sources, depending design amp. common source thermal feedback. temperature shift sole cause nonlinearity error, assumed that minimizing output loading will help. verify this, nonlinearity measured with load then compared loaded condition. oscilloscope display test circuit measuring open loop gain nonlinearity shown Figure 3.7. same precautions previously discussed relating offset voltage test circuit must observed this circuit. amplifier configured signal gain open loop gain defined change output voltage divided change input offset voltage. However, large values AVOL, offset change only microvolts over entire output voltage swing. Therefore divider consisting resistor forces voltage
AMPLIFIERS SIGNAL CONDITIONING
100,001 value chosen give measurable voltages depending expected values VOS.
CIRCUIT MEASURES OPEN LOOP GAIN NONLINEARITY
-VREF (-10V) +VREF (+10V) AVOL -15V NONLINEAR +15V IDEAL
±10V RAMP
OFFSET ADJUST (Multi-Turn Film-Type)
CLOSED LOOP GAIN NONLINEARITY
OPEN LOOP GAIN NONLINEARITY
AVOL,MIN AVOL,MAX
Figure ±10V ramp generator output multiplied signal gain, forces output voltage swing from +10V -10V. Because gain factor applied offset voltage, offset adjust potentiometer added allow initial output offset zero. resistor values chosen will null input offset voltage ±10mV. Stable voltage references (AD688) should used each potentiometer prevent output drift. Also, frequency ramp generator must quite low, probably more than fraction because corner frequency open loop gain (0.1Hz OP177). plot right-hand side Figure shows plotted against there gain nonlinearity graph will have constant slope, AVOL calculated follows: 100,001
there nonlinearity, AVOL will vary output signal changes. approximate open loop gain nonlinearity calculated based maximum minimum values AVOL over output voltage range:
AMPLIFIERS SIGNAL CONDITIONING
Open Loop Gain Nonlinearity
VOL,MIN
VOL,MAX
closed loop gain nonlinearity obtained multiplying open loop gain nonlinearity noise gain, Closed Loop Gain Nonlinearity VOL,MIN VOL,MAX ideal case, plot versus would have constant slope, reciprocal slope open loop gain, AVOL. horizontal line with zero slope would indicate infinite open loop gain. actual amp, slope change across output range because nonlinearity, thermal feedback, etc. fact, slope even change sign. Figure shows (and VOS) versus plot OP177 precision amp. plot shown different loads, 10k. reciprocal slope calculated based points, average AVOL about million. maximum minimum values AVOL across output voltage range measured approximately million, million, respectively. This corresponds open loop gain nonlinearity about 0.07ppm. Thus, noise gain 100, corresponding closed loop gain nonlinearity about 7ppm.
OP177 GAIN NONLINEARITY
50mV DIV.
(0.5µV DIV.) (RTI)
AVOL
-10V
AVOL,MAX million, AVOL,MIN 5.7million OPEN LOOP GAIN NONLINEARITY 0.07ppm CLOSED LOOP GAIN NONLINEARITY Figure
AVOL (AVERAGE) million
OUTPUT VOLTAGE
+10V
Noise
three noise sources circuit voltage noise amp, current noise (there uncorrelated sources, each input), 3.10
AMPLIFIERS SIGNAL CONDITIONING Johnson noise resistances circuit. noise components "white" noise medium frequencies frequency "1/f" noise, whose spectral density inversely proportional square root frequency. should noted that, though both voltage current noise have same characteristic behavior, particular amplifier corner frequency necessarily same voltage current noise usually specified voltage noise shown Figure 3.9.
INPUT VOLTAGE NOISE OP177/AD707
INPUT VOLTAGE NOISE, TIME 1sec/DIV.
0.1Hz 10Hz VOLTAGE NOISE
CORNER 0.7Hz (WHITE)
200nV
FREQUENCY (Hz) Vn,rms 0.1Hz, 10Hz, 10nV/ 0.7Hz: Vn,rms 36nV Vn,pp 36nV 238nV
Figure frequency noise generally known noise (the noise power obeys noise voltage noise current proportional 1/f). frequency which noise spectral density equals white noise known corner frequency, figure merit amp, with corner frequencies indicating better performance. Values corner frequency vary from less than high accuracy bipolar amps like OP177/AD707, several hundred AD743/745 FET-input amps, several thousands some high speed amps where process compromises favor high speed rather than frequency noise. OP177/AD707 shown Figure 3.9, corner frequency 0.7Hz, white noise 10nV/Hz. frequency noise often expressed peak-to-peak noise bandwidth 0.1Hz 10Hz shown scope photo Figure 3.9. Note that this noise ultimately limits resolution precision measurement system because bandwidth 10Hz usually bandwidth most interest. equation total noise, Vn,rms, bandwidth given equation:
3.11
AMPLIFIERS SIGNAL CONDITIONING
Vn,rms where noise spectral density "white noise" region (usually specified frequency 1kHz), corner frequency, measurement bandwidth interest. example shown, 0.1Hz 10Hz noise calculated 36nV rms, approximately 238nV peak-to-peak, which closely agrees with scope photo right factor generally used convert values peak-to-peak values). should noted that higher frequencies, term equation containing natural logarithm becomes insignificant, expression noise becomes:
Vn,rms And, Vn,rms However, some amps (such OP07 OP27) have voltage noise characteristics that increase slightly high frequencies. voltage noise versus frequency curve amps should therefore examined carefully flatness when calculating high frequency noise using this approximation. very frequencies when operating exclusively region, -FL), expression noise reduces Vn,rms Note that there reducing this noise filtering operation extends Making FH=0.1Hz 0.001 still yields noise about 18nV rms, 119nV peak-to-peak. point that averaging results large number measurements taken over long period time practically effect error produced noise. only method reducing further chopper stabilized which does pass frequency noise components. generalized noise model shown Figure 3.10. uncorrelated noise sources root-sum-of-squares manner, i.e., noise voltages give result
3.12
AMPLIFIERS SIGNAL CONDITIONING Thus, noise voltage which more than times others dominant, others generally ignored. This simplifies noise analysis. this diagram, total noise sources shown referred input (RTI). noise useful because compared directly input signal level. total noise referred output (RTO) obtained simply multiplying noise noise gain. diagram assumes that feedback network purely resistive. contains reactive elements (usually capacitors), noise gain constant over bandwidth interest, more complex techniques must used calculate total noise (see particular, Reference 12). However, precision applications where feedback network most likely resistive, equations valid. Notice that Johnson noise voltage associated with three resistors been included. resistors have Johnson noise 4kTBR where Boltzmann's Constant J/K), absolute temperature, bandwidth resistance simple relationship which easy remember that 1000 resistor generates Johnson noise 4nV/Hz
NOISE MODEL
VN,R2
VN,R1
GAIN FROM OUTPUT NOISE GAIN
4kTR1 VN,R3
4kTR2
CLOSED LOOP
VOUT
GAIN FROM OUTPUT
4kTR3
4kTR3 R1+R2
4kTR1 R1+R2 4kTR2 R1+R2
NOISE
IN+2R32 IN-2
NOISE NOISE
1.57
Figure 3.10
3.13
AMPLIFIERS SIGNAL CONDITIONING voltage noise various amps vary from under 1nV/Hz 20nV/Hz, even more. Bipolar input amps tend have lower voltage noise than JFET input ones, although possible make JFET input amps with voltage noise (such AD743/AD745), cost large input devices hence large (~20pF) input capacitance. Current noise vary much more widely, from around 0.1fA/Hz JFET input electrometer amps) several pA/Hz high speed bipolar amps). bipolar JFET input devices where bias current flows into input junction, current noise simply Schottky shot) noise bias current. shot noise spectral density simply amps/Hz, where bias current amps) charge electron cannot calculated bias-compensated current feedback amps where external bias current difference between internal current sources. Current noise only important when flows through impedance turn generates noise voltage. equation shown Figure 3.10 shows current noise flowing resistors contribute total noise. choice noise therefore depends impedances around Consider OP27, bias compensated with voltage noise (3nV/Hz), quite high current noise (1pA/Hz) shown schematic Figure 3.11. With zero source impedance, voltage noise dominates. With source resistance current noise (1pA/Hz) flowing will equal voltage noise, Johnson noise resistor 7nV/Hz dominant. With source resistance 300k, effect current noise increases hundredfold 300nV/Hz, while voltage noise unchanged, Johnson noise (which proportional square root resistance) increases tenfold. Here, current noise dominates.
DIFFERENT NOISE SOURCES DOMINATE DIFFERENT SOURCE IMPEDANCES
EXAMPLE: OP27 Voltage Noise Current Noise 25°C
OP27 Neglect Noise Contribution JOHNSON NOISE CONTRIBUTION FROM AMPLIFIER VOLTAGE NOISE AMPLIFIER CURRENT NOISE FLOWING VALUES 300k
NOISE Dominant Noise Source Highlighted
Figure 3.11
3.14
AMPLIFIERS SIGNAL CONDITIONING above example shows that choice noise depends source impedance input signal, high impedances, current noise always dominates. This shown Figure 3.12 several bipolar (OP07, OP27, 741) JFET (AD645, AD743, AD744) amps. impedance circuitry (generally 1k), amplifiers with voltage noise, such OP27 will obvious choice, their comparatively large current noise will affect application. medium resistances, Johnson noise resistors dominant, while very high resistances, must choose with smallest possible current noise, such AD549 AD645. Until recently, BiFET amplifiers (with JFET inputs) tended have comparatively high voltage noise (though very current noise), thus were more suitable noise applications high rather than impedance circuitry. AD645, AD743, AD745 have very values both voltage current noise. AD645 specifications 10kHz 10nV/Hz 0.6fA/Hz, AD743/AD745 specifications 10kHz 2.0nV/Hz 6.9fA/Hz. These make possible design noise amplifier circuits which have noise over wide range source impedances.
DIFFERENT AMPLIFIERS BEST DIFFERENT SOURCE IMPEDANCE LEVELS
OP27,
OP07
OP27 OP07,
Vertical Scales
OP27 OP07
Horizontal Scales
Figure 3.12
3.15
AMPLIFIERS SIGNAL CONDITIONING
Common Mode Rejection Power Supply Rejection
signal applied equally both inputs that differential input voltage unaffected, output should affected. practice, changes common mode voltage will produce changes output. common mode rejection ratio CMRR ratio common mode gain differentialmode gain amp. example, differential input change volts will produce change output, common mode change volts produces similar change then CMRR X/Y. normally expressed typical values between 120dB. When expressed generally referred common mode rejection (CMR). higher frequencies, deteriorates many data sheets show plot versus frequency shown Figure 3.13 OP177/AD707 precision amps. CMRR produces corresponding output offset voltage error amps configured non-inverting mode shown Figure 3.14. amps configured inverting mode have CMRR output error because both inputs ground virtual ground, there common mode voltage, only offset voltage amplifier un-nulled.
OP177/AD707 COMMON MODE REJECTION (CMR)
0.01 100k log10 CMRR
FREQUENCY
Figure 3.13
3.16
AMPLIFIERS SIGNAL CONDITIONING
CALCULATING OFFSET ERROR COMMON MODE REJECTION RATIO (CMRR)
VOUT
ERROR (RTI)
CMRR
CMRR
VOUT
CMRR CMRR
ERROR (RTO)
Figure 3.14 supply changes, output should not, will. specification power supply rejection ratio PSRR defined similarly definition CMRR. change volts supply produces same output change differential input change volts, then PSRR that supply X/Y. When ratio expressed generally referred power supply rejection, PSR. definition PSRR assumes that both supplies altered equally opposite directions otherwise change will introduce common mode change well supply change, analysis becomes considerably more complex. this effect which causes apparent differences PSRR between positive negative supplies. case single supply amps, generally defined with respect change positive supply. Many single supply amps have separate specifications positive negative supplies. OP177/AD707 shown Figure 3.15. PSRR amps frequency dependent, therefore power supplies must well decoupled shown Figure 3.16. frequencies, several devices share 50µF capacitor each supply, provided more than 10cm track distance) from them. high frequencies, each must have every supply decoupled inductance capacitor (0.1µF with short leads tracks. These capacitors must also provide return path currents load. Decoupling capacitors should connected impedance large area ground plane with minimum lead lengths. Surface mount capacitors minimize lead inductance good choice.
3.17
AMPLIFIERS SIGNAL CONDITIONING
OP177/AD707 POWER SUPPLY REJECTION (PSR)
0.01 100k log10 PSRR
FREQUENCY
Figure 3.15
PROPER HIGH-FREQUENCY DECOUPLING TECHNIQUES AMPS
10cm LARGE AREA GROUND PLANE
LEAD LENGTH MINIMUM
10cm
LOCALIZED DECOUPLING, INDUCTANCE CERAMIC, 0.1µF
SHARED DECOUPLING, ELECTROLYTIC, 50µF
Figure 3.16
3.18
AMPLIFIERS SIGNAL CONDITIONING
AMPLIFIER ERROR BUDGET ANALYSIS
room temperature error budget analysis OP177A shown Figure 3.17. amplifier connected inverting mode with signal gain 100. data sheet specifications also shown diagram. assume input signal 100mV fullscale which corresponds output signal 10V. various error sources normalized fullscale expressed parts million (ppm). Note: parts million (ppm) error fractional error error 104. Note that offset errors gain error finite AVOL removed with system calibration. However, error open loop gain nonlinearity cannot removed with calibration produces relative accuracy error, often called resolution error. second contributor resolution error noise. This noise always present adds uncertainty measurement. overall relative accuracy circuit room temperature 9ppm which equivalent approximately bits resolution.
PRECISION (OP177A) ERROR BUDGET
OP177A AVOL AVOL Nonlinearity 0.1Hz 10Hz Noise Total Unadjusted Error Resolution Error VOUT 10µV 100mV 100mV (100/ 100mV 0.07ppm 100ppm 1ppm 20ppm 7ppm MAXIMUM ERROR CONTRIBUTION, 25°C FULLSCALE: VIN=100mV, VOUT
200nV 100mV
2ppm
SPECS +25°C: 10µV AVOL AVOL Nonlinearity 0.07ppm 0.1Hz 10Hz Noise 200nV
Bits Accurate Bits Accurate
130ppm
9ppm
Figure 3.17
3.19
AMPLIFIERS SIGNAL CONDITIONING
SINGLE SUPPLY AMPS
Over last several years, single-supply operation become increasingly important requirement because market requirements. Automotive, set-top box, camera/cam-corder, laptop computer applications demanding vendors supply array linear devices that operate single supply rail, with same performance dual supply parts. Power consumption parameter line battery operated systems, some instances, more important than cost. This makes low-voltage/low supply current operation critical; same time, however, accuracy precision requirements have forced manufacturers meet challenge "doing more with less" their amplifier designs.
SINGLE SUPPLY AMPLIFIERS
Single Supply Offers: Lower Power Battery Operated Portable Equipment Requires Only Voltage Design Tradeoffs: Reduced Signal Swing Increases Sensitivity Errors Caused Offset Voltage, Bias Current, Finite OpenLoop Gain, Noise, etc. Must Usually Share Noisy Digital Supply Rail-to-Rail Input Output Needed Increase Signal Swing Precision Less than best Dual Supply Amps Required Applications Many Amps Specified Single Supply, have Rail-to-Rail Inputs Outputs
Figure 3.18 single-supply application, most immediate effect performance amplifier reduced input output signal range. result these lower input output signal excursions, amplifier circuits become more sensitive internal external error sources. Precision amplifier offset voltages order 0.1mV less than 0.04 error source 12-bit, full-scale system. single-supply system, however, "rail-to-rail" precision amplifier with offset voltage represents 0.8LSB error fullscale system, 1.6LSB error 2.5V fullscale system.
3.20
AMPLIFIERS SIGNAL CONDITIONING keep battery current drain low, larger resistors usually used around amp. Since bias current flows through these larger resistors, they generate offset errors equal greater than amplifier's offset voltage. Gain accuracy some voltage single-supply devices also reduced, device selection needs careful consideration. Many amplifiers having open-loop gains millions typically operate dual supplies: example, OP07 family types. However, many single-supply/rail-to-rail amplifiers precision applications typically have open-loop gains between 25,000 30,000 under light loading (>10k). Selected devices, like OP113/213/413 family, have high open-loop gains (i.e., 1M). Many trade-offs possible design single-supply amplifier circuit: speed versus power, noise versus power, precision versus speed power, etc. Even noise floor remains constant (highly unlikely), signal-to-noise ratio will drop signal amplitude decreases. Besides these limitations, many other design considerations that otherwise minor issues dual-supply amplifiers become important. example, signalto-noise (SNR) performance degrades result reduced signal swing. "Ground reference" longer simple choice, reference voltage work some devices, others. Amplifier voltage noise increases operating supply current drops, bandwidth decreases. Achieving adequate bandwidth required precision with somewhat limited selection amplifiers presents significant system design challenges single-supply, low-power applications. Most circuit designers take "ground" reference granted. Many analog circuits scale their input output ranges about ground reference. dual-supply applications, reference that splits supplies (0V) very convenient, there equal supply headroom each direction, generally voltage impedance ground plane. single-supply/rail-to-rail circuits, however, ground reference chosen anywhere within supply range circuit, since there standard follow. choice ground reference depends type signals processed amplifier characteristics. example, choosing negative rail ground reference optimize dynamic range whose output designed swing other hand, signal require level shifting order compatible with input other devices (such ADCs) that designed operate input. Early single-supply "zero-in, zero-out" amplifiers were designed bipolar processes which optimized performance transistors. transistors were either lateral substrate PNPs with much less bandwidth than NPNs. Fully complementary processes required new-breed single-supply/railto-rail operational amplifiers. These amplifier designs lateral substrate transistors within signal path, incorporate parallel input stages accommodate input signal swings from ground positive supply rail. Furthermore, rail-to-rail output stages designed with bipolar common-emitter, N-channel/P-channel common-source amplifiers whose
3.21
AMPLIFIERS SIGNAL CONDITIONING collector-emitter saturation voltage drain-source channel on-resistance determine output signal swing function load current. characteristics single-supply amplifier input stage (common mode rejection, input offset voltage temperature coefficient, noise) critical precision, low-voltage applications. Rail-to-rail input operational amplifiers must resolve small signals, whether their inputs ground, some cases near amplifier's positive supply. Amplifiers having minimum 60dB common mode rejection over entire input common mode voltage range from positive supply good candidates. necessary that amplifiers maintain common mode rejection signals beyond supply voltages: what required that they self-destruct momentary overvoltage conditions. Furthermore, amplifiers that have offset voltages less than offset voltage drifts less than 2µV/°C also very good candidates precision applications. Since input signal dynamic range equally more important than output dynamic range SNR, precision single-supply/rail-to-rail operational amplifiers should have noise levels referred-to-input (RTI) less than 5µVp-p 0.1Hz 10Hz band. need rail-to-rail amplifier output stages driven need maintain wide dynamic range low-supply voltage applications. single-supply/rail-to-rail amplifier should have output voltage swings which within least 100mV either supply rail (under nominal load). output voltage swing very dependent output stage topology load current. voltage swing good output stage should maintain rated swing loads down 10k. smaller larger VOH, better. System parameters, such "zeroscale" "full-scale" output voltage, should determined amplifier's (for zero-scale) (for full-scale). Since majority single-supply data acquisition systems require least 14-bit performance, amplifiers which exhibit open-loop gain greater than 30,000 loading conditions good choices precision applications.
Single Supply Input Stages
There some demand amps whose input common mode voltage includes both supply rails. Such feature undoubtedly useful some applications, engineers should recognize that there relatively applications where absolutely essential. These should carefully distinguished from many applications where common mode range close supplies that includes supplies necessary, input rail-to-rail operation not. many single-supply applications, required that input only supply rails (usually ground). High-side low-side sensing applications good examples this. Amplifiers which will handle zero-volt inputs relatively easily designed using differential pairs N-channel JFET pairs) shown Figure 3.19. input common mode range such extends from about 200mV below negative supply within about positive supply.
3.22
AMPLIFIERS SIGNAL CONDITIONING
N-CHANNEL JFET STAGES ALLOW INPUT SIGNAL NEGATIVE RAIL
PNPs
N-CH JFETs
Figure 3.19 input stage could also designed with transistors P-channel JFETs), which case input common mode range would include positive rail within about negative rail. This requirement typically occurs applications such high-side current sensing, low-frequency measurement application. OP282/OP482 input stage uses P-channel JFET input pair whose input common mode range includes positive rail. Other circuit topologies high-side sensing (such AD626) precision resistors attenuate common mode voltage. True rail-to-rail input stages require long-tailed pairs (see Figure 3.20), bipolar transistors N-channel JFETs), other transistors P-channel JFETs). These pairs exhibit different offsets bias currents, when applied input common mode voltage changes, amplifier input offset voltage input bias current does also. fact, when both current sources remain active throughout entire input common mode range, amplifier input offset voltage average offset voltage pair pair. those designs where current sources alternatively switched some point along input common mode voltage, amplifier input offset voltage dominated pair offset voltage signals near negative supply, pair offset voltage signals near positive supply. should noted that true railto-rail input stages also constructed from CMOS transistors case OP250/450 AD8531/8532/8534.
3.23
AMPLIFIERS SIGNAL CONDITIONING
TRUE RAIL-TO-RAIL INPUT STAGE
Figure 3.20 Amplifier input bias current, function transistor current gain, also function applied input common mode voltage. result relatively poor common mode rejection (CMR), changing common mode input impedance over common mode input voltage range, compared familiar dual-supply devices. These specifications should considered carefully when choosing rail-rail input amp, especially non-inverting configuration. Input offset voltage, input bias current, even quite good over part common mode range, much worse region where operation shifts between devices vice versa. True rail-to-rail amplifier input stage designs must transition from differential pair other differential pair somewhere along input common mode voltage range. Some devices like OP191/291/491 family OP279 have common mode crossover threshold approximately below positive supply. differential input stage active from about 200mV below negative supply within about positive supply. Over this common mode range, amplifier input offset voltage, input bias current, CMR, input noise voltage/current primarily determined characteristics differential pair. crossover threshold, however, amplifier input offset voltage becomes average offset voltage NPN/PNP pairs change rapidly. Also, amplifier bias currents, dominated differential pair over most input common mode range, change polarity magnitude crossover threshold when differential pair becomes active. amps like OP184/284/484, utilize rail-to-rail input stage design where both transistor pairs active throughout entire input common mode voltage range, there common mode crossover threshold. Amplifier input offset voltage average offset voltage stages. Amplifier 3.24
AMPLIFIERS SIGNAL CONDITIONING input offset voltage exhibits smooth transition throughout entire input common mode range because careful laser trimming resistors input stage. same manner, through careful input stage current balancing input transistor design, amplifier input bias currents also exhibit smooth transition throughout entire common mode input voltage range. exception occurs extremes input common mode range, where amplifier offset voltages bias currents increase sharply slight forward-biasing parasitic junctions. This occurs input voltages within approximately either supply rail. When both differential pairs active throughout entire input common mode range, amplifier transient response faster through middle common mode range much factor bipolar input stages factor JFET input stages. Input stage transconductance determines slew rate unity-gain crossover frequency amplifier, hence response time degrades slightly extremes input common mode range when either stage (signals approaching positive supply rail) stage (signals approaching negative supply rail) forced into cutoff. thresholds which transconductance changes occur approximately within either supply rail, behavior similar that input bias currents. Applications which require true rail-rail inputs should therefore carefully evaluated, amplifier chosen ensure that input offset voltage, input bias current, common mode rejection, noise (voltage current) suitable.
Single Supply Output Stages
earliest output stages were emitter followers with current sources resistive pull-downs, shown left-hand diagram Figure 3.21. Naturally, slew rates were greater positive-going than negative-going signals. While modern amps have push-pull output stages some sort, many still asymmetrical, have greater slew rate direction than other. Asymmetry tends introduce distortion signals generally results from processes with faster than transistors. also result ability output approach supply more closely than other. many applications, output required swing only rail, usually negative rail (i.e., ground single-supply systems). pulldown resistor negative rail will allow output approach that rail (provided load impedance high enough, also grounded that rail), only slowly. Using current source instead resistor speed things this adds complexity. With complementary bipolar processes (CB), well matched high speed transistors available. complementary emitter follower output stage shown right-hand diagram Figure 3.21 many advantages including output impedance. However, output only swing within about drop either supply rail. output swing typical such stages when operated single supply.
3.25
AMPLIFIERS SIGNAL CONDITIONING
TRADITIONAL OUTPUT STAGES
VOUT
NMOS VOUT
VOUT
NMOS
Figure 3.21 complementary common-emitter/common-source output stages shown Figure 3.22 allow output voltage swing much closer output rails, these stages have higher open loop output impedance than emitter follower- based stages. practice, however, amplifier's open loop gain local feedback produce apparent output impedance, particularly frequencies below 10Hz. complementary common emitter output stage using BJTs (left-hand diagram Figure 3.22) cannot swing completely rails, only within transistor saturation voltage (VCESAT) rails. small amounts load current (less than 100µA), saturation voltage 10mV, higher load currents, saturation voltage increase several hundred (for example, 500mV 50mA). other hand, output stage constructed CMOS FETs provide nearly true rail-to-rail performance, only under no-load conditions. output must source sink current, output swing reduced voltage dropped across FETs internal "on" resistance (typically, precision amplifiers, less than high current drive CMOS amplifiers). these reasons, apparent that there such thing true rail-to-rail output stage, hence title Figure 3.22 ("Almost" Rail-to-Rail Output Stages).
3.26
AMPLIFIERS SIGNAL CONDITIONING
"ALMOST" RAIL-TO-RAIL OUTPUT STRUCTURES
VOUT
PMOS VOUT
NMOS
SWINGS LIMITED SATURATION VOLTAGE
SWINGS LIMITED "ON" RESISTANCE
Figure 3.22 Figure 3.23 summarizes performance characteristics number singlesupply amps suitable some precision applications. devices listed order increasing supply current. Single, dual, quad versions each available, supply current normalized ISY/amplifier comparison. input output voltage ranges +5V) also supplied table. inputs pairs, with exception AD820/822/824 which NChannel JFETs. Output stages having voltage ranges designated "5mV, emitter-followers with current source pull-downs (OP193/293/493, OP113/213/413). Output stages designated "R/R" CMOS common source stages (OP181/281/481) common emitter stages (OP196/296/496, OP191/291/491, AD820/822/824, OP184/284/484). summary, following points should considered when selecting amplifiers single-supply/rail-to-rail applications: First, input offset voltage input bias currents function applied input common mode voltage (for true rail-to-rail input amps). Circuits using this class amplifiers should designed minimize resulting errors. inverting amplifier configuration with false ground reference non-inverting input prevents these errors holding input common mode voltage constant. inverting amplifier configuration cannot used, then amplifiers like OP184/284/OP484 which exhibit common mode crossover thresholds should used.
3.27
AMPLIFIERS SIGNAL CONDITIONING
PRECISION SINGLE-SUPPLY PERFORMANCE CHARACTERISTICS
**LISTED ORDER INCREASING SUPPLY CURRENT **PART OP181/281/481 OP193/293/493 OP196/296/496 OP191/291/491 *AD820/822/824 OP184/284/484 OP113/213/413 1500µV 75µV 300µV 700µV 400µV 65µV 125µV AVOLmin NOISE (1kHz) INPUT OUTPUT ISY/AMP 10µV/°C 0.2µV/°C 1.5µV/°C 1.1µV/°C 2µV/°C 0.2µV/°C 0.2µV/°C 200k 150k 500k 70nV/Hz 65nV/Hz 26nV/Hz 35nV/Hz 16nV/Hz 3.9nV/Hz 4.7nV/Hz "R/R" 5mV, "R/R" "R/R" "R/R" "R/R" 15µA 50µA 400µA 800µA 1250µA
5mV, 1750µA
*JFET INPUT
NOTE: Unless Otherwise Stated Specifications Typical +25°C
Figure 3.23 Second, since input bias currents always small exhibit different polarities, source impedance levels should carefully matched minimize additional input bias current-induced offset voltages increased distortion. Again, consider using amplifiers that exhibit smooth input bias current transition throughout applied input common mode voltage. Third, rail-to-rail amplifier output stages exhibit load-dependent gain which affects amplifier open-loop gain, hence closed-loop gain accuracy. Amplifiers with openloop gains greater than 30,000 resistive loads less than good choices precision applications. applications requiring full rail-rail swings, device families like OP113/213/413 OP193/293/493 offer gains 200,000 more. Lastly, matter what claims made, rail-to-rail output voltage swings functions amplifier's output stage devices load current. saturation voltage (VCESAT), saturation resistance (RSAT) bipolar output stages, on-resistance CMOS output stages, well load current affect amplifier output voltage swing.
Process Technologies
wide variety processes used make amps shown Figure 3.24. earliest amps were made using standard NPN-based bipolar processes. transistors available these processes were extremely slow were used primarily current sources level shifting. 3.28
AMPLIFIERS SIGNAL CONDITIONING
ability produce matching high speed transistors bipolar process added great flexibility circuit designs. These complementary bipolar (CB) processes widely used today's precision amps, well those requiring wide bandwidths. high-speed transistors have which greater than one-half NPNs. addition JFETs complementary bipolar process (CBFET) allow high input impedance amps designed suitable such applications photodiode electrometer preamplifiers. CMOS amps, with exceptions, generally have relatively poor offset voltage, drift, voltage noise. However, input bias current very low. They offer power cost, however, improved performance achieved with BiFET CBFET processes. addition bipolar complementary devices CMOS process (BiMOS CBCMOS) adds great flexibility, better linearity, power. bipolar devices typically used input stage provide good gain linearity, CMOS devices rail-to-rail output stage. summary, there single process which optimum amps. Process selection resulting design depends targeted applications ultimately should transparent customer.
PROCESS TECHNOLOGY SUMMARY
BIPOLAR (NPN-BASED): This Where Started!! COMPLEMENTARY BIPOLAR (CB): Rail-to-Rail, Precision, High Speed BIPOLAR JFET (BiFET): High Input Impedance, High Speed COMPLEMENTARY BIPOLAR JFET (CBFET): High Input Impedance, Rail-to-Rail Output, High Speed COMPLEMENTARY MOSFET (CMOS): Cost, Non-Critical Amps BIPOLAR CMOS (BiCMOS): Bipolar Input Stage adds Linearity, Power, Rail-to-Rail Output COMPLEMENTARY BIPOLAR CMOS (CBCMOS): Rail-to-Rail Inputs, Rail-to-Rail Outputs, Good Linearity, Power Figure 3.24
3.29
AMPLIFIERS SIGNAL CONDITIONING
INSTRUMENTATION AMPLIFIERS (IN-AMPS)
instrumentation amplifier closed-loop gain block which differential input output which single-ended with respect reference terminal (see Figure 3.25). input impedances balanced have high values, typically higher. Unlike amp, which closed-loop gain determined external resistors connected between inverting input output, in-amp employs internal feedback resistor network which isolated from signal input terminals. With input signal applied across differential inputs, gain either preset internally user-set internal (via pins) external gain resistor, which also isolated from signal inputs. Typical in-amp gain settings range from 10,000.
INSTRUMENTATION AMPLIFIER
RS/2
COMMON MODE VOLTAGE
VSIG
IN-AMP GAIN VREF VOUT
VSIG
RS/2
CMRR
COMMON MODE ERROR (RTI) Figure 3.25
order effective, in-amp needs able amplify microvolt-level signals, while simultaneously rejecting volts common mode signal inputs. This requires that in-amps have very high common mode rejection (CMR): typical values 70dB over 100dB, with usually improving higher gains. important note that specification inputs alone sufficient most practical applications. industrial applications, most common cause external interference pickup from 50/60Hz power mains. Harmonics power mains frequency also troublesome. differential measurements, this type interference tends induced equally onto both in-amp inputs. interfering signal therefore appears common mode signal in-amp. Specifying over frequency more important than specifying value. Imbalance source impedance degrade some in-amps. Analog Devices fully specifies in-amp 50/60Hz with source impedance imbalance
3.30
AMPLIFIERS SIGNAL CONDITIONING Low-frequency amps, connected subtractors shown Figure 3.26, generally function resistors around circuit, amp. mismatch only 0.1% resistor ratios will reduce approximately 66dB. Another problem with simple subtractor that input impedances relatively unbalanced between sides. input impedance seen input impedance seen R2'. This configuration quite problematic terms CMR, since even small source impedance imbalance will degrade workable CMR.
SUBTRACTOR
VOUT Where Total Fractional Mismatch log10
VOUT
CRITICAL HIGH EXTREMELY SENSITIVE SOURCE IMPEDANCE IMBALANCE 0.1% TOTAL MISMATCH YIELDS Figure 3.26
66dB
Instrumentation Amplifier Configurations
Instrumentation amplifier configurations based amps, simple subtractor circuit described above lacks performance required precision applications. in-amp architecture which overcomes some weaknesses subtractor circuit uses amps shown Figure 3.27. This circuit typically referred in-amp. Dual amps used most cases good matching. circuit gain trimmed with external resistor, input impedance high, permitting impedance signal sources high unbalanced. common mode rejection limited matching R1/R2 R1'/R2'. there mismatch four resistors, common mode rejection limited GAIN %MISMATCH
3.31
AMPLIFIERS SIGNAL CONDITIONING
INSTRUMENTATION AMPLIFIER
VREF VOUT
VREF
VOUT 20log GAIN MISMATCH
Figure 3.27 There implicit advantage this configuration gain executed signal. This raises proportion. Integrated instrumentation amplifiers particularly well suited meeting combined needs ratio matching temperature tracking gain-setting resistors. While thin film resistors fabricated silicon have initial tolerance ±20%, laser trimming during production allows ratio error between resistors reduced 0.01% (100ppm). Furthermore, tracking between temperature coefficients thin film resistors inherently typically less than When dual supplies used, VREF normally connected directly ground. single supply applications, VREF usually connected impedance voltage source equal one-half supply voltage. gain from VREF node R1/R2, gain from node output R2'/R1'. This makes gain from VREF output equal unity, assuming perfect ratio matching. Note that critical that source impedance seen VREF low, otherwise will degraded. major disadvantage this design that common mode voltage input range must traded against gain. amplifier must amplify signal
3.32
AMPLIFIERS SIGNAL CONDITIONING (low gain Figure 3.27), will saturate common mode signal high, leaving headroom amplify wanted differential signal. high gains (R1<< R2), there correspondingly more headroom node allowing larger common mode input voltages. common mode rejection this configuration generally poor because signal from VOUT additional phase shift addition, amplifiers operating different closed-loop gains (and thus different bandwidths). small trim capacitor shown diagram improve somewhat. gain single supply in-amp configuration results when used, shown Figure 3.28. input common mode differential signals must limited values which prevent saturation either example, amps remain linear within 0.1V supply rails, their upper lower output limits designated VOL, respectively. Using equations shown diagram, voltage must fall between 1.3V 2.4V prevent from saturating. Notice that VREF connected average (2.5V). This allows bipolar differential input signals with VOUT referenced +2.5V.
SINGLE SUPPLY RESTRICTIONS: +5V,
VREF 2.5V V1,MAX 1)VOH VREF 3.7V 2.4V VOH=4.9V VOL=0.1V VOUT VOH=4.9V VOL=0.1V
V1,MIN
1.3V
VREF 2.5V
Figure 3.28 high gain 100) single supply in-amp configuration shown Figure 3.29. Using same equations, note that voltage swing between 0.124V 4.876V. Again, VREF connected 2.5V allow bipolar differential input output signals.
3.33
AMPLIFIERS SIGNAL CONDITIONING
SINGLE SUPPLY RESTRICTIONS: +5V,
990k VREF 2.5V V1,MAX 1)VOH VREF 4.876V 0.048V VOH=4.9V VOL=0.1V VOUT VOH=4.9V VOL=0.1V 990k
V1,MIN
0.124V
VREF 2.5V
Figure 3.29
above discussion shows that regardless gain, basic in-amp does allow zero-volt common mode input voltages when operated single supply. This limitation overcome using circuit shown Figure 3.30 which implemented AD627 in-amp. Each composed common emitter input stage gain stage, designated Q1/A1 Q2/A2, respectively. transistors only provide gain also level shift input signal positive about 0.5V, thereby allowing common mode input voltage 0.1V below negative supply rail. maximum positive input voltage allowed less than positive supply rail. AD627 in-amp delivers rail-to-rail output swing operates over wide supply voltage range (+2.7V ±18V). Without external gain setting resistor, in-amp gain Gains 1000 with single external resistor. Common mode rejection AD627B 60Hz with source imbalance 85dB when operating single supply Even though AD627 in-amp, patented circuit keeps flat much higher frequency than would achievable with conventional discrete in-amp. AD627 data sheet (available http://www.analog.com) detailed discussion allowable input/output voltage ranges function gain power supply voltages. specifications AD627 summarized Figure 3.31.
3.34
AMPLIFIERS SIGNAL CONDITIONING
AD627 IN-AMP ARCHITECTURE
100k
VOUT
100k
200k
VOUT G(V2 VREF
VREF
Figure 3.30
AD627 IN-AMP SPECIFICATIONS
Wide Supply Range +2.7V ±18V Input Voltage Range: 0.1V 85µA Supply Current Gain Range: 1000 75µV Maximum Input Offset Volage (AD627B) 10ppm/°C Maximum Offset Voltage (AD627B) 10ppm Gain Nonlinearity 85dB 60Hz, Source Imbalance 0.1Hz 10Hz Input Voltage Noise Figure 3.31 true balanced high impedance inputs, three amps connected form in-amp shown Figure 3.32. This circuit typically referred three in-amp. gain amplifier resistor, which internal, external, (software pin-strap) programmable. this configuration, depends upon ratio matching R3/R2 R3'/R2'. Furthermore, common mode signals only amplified factor regardless gain common mode voltage will appear across hence, common mode current will flow 3.35
AMPLIFIERS SIGNAL CONDITIONING because input terminals will have significant potential difference between them). Thus, will theoretically increase direct proportion gain. Large common mode signals (within A1-A2 headroom limits) handled gains. Finally, because symmetry this configuration, common mode errors input amplifiers, they track, tend canceled subtractor output stage. These features explain popularity three in-amp configuration.
THREE INSTRUMENTATION AMPLIFIER
VSIG
VOUT
VSIG
VOUT VSIG
VREF
VREF
20log
GAIN MISMATCH
Figure 3.32 classic three configuration been used number monolithic instrumentation amplifiers. Besides offering excellent matching between three internal amps, thin film laser trimmed resistors provide excellent ratio matching gain accuracy much lower cost than using discrete amps resistor networks. AD620 excellent example monolithic in-amp technology, simplified schematic shown Figure 3.33. AD620 highly popular in-amp specified power supply voltages from ±2.3V ±18V. Input voltage noise only 9nV/Hz 1kHz. Maximum input bias current only maximum because Superbeta input stage. Overvoltage protection provided internal thin-film current-limit resistors conjunction with diodes which connected from emitter-tobase gain with single external resistor. appropriate internal resistors trimmed that standard 0.1% resistors used AD620 gain popular gain values.
3.36
AMPLIFIERS SIGNAL CONDITIONING
AD620 IN-AMP SIMPLIFIED SCHEMATIC
49.4k
24.7k 24.7k
VREF
Figure 3.33
case in-amp configuration, single supply operation three in-amp requires understanding internal node voltages. Figure 3.34 shows generalized diagram in-amp operating single supply. maximum minimum allowable output voltages individual amps designated (maximum high output) (minimum output) respectively. Note that gain from common mode voltage outputs unity, that common mode voltage signal voltage these outputs must fall within amplifier output voltage range. obvious that this configuration cannot handle input common mode voltages either zero volts because saturation case in-amp, output reference positioned halfway between order allow bipolar differential input signals. This chapter emphasized operation high performance linear circuits from single, low-voltage supply less) common requirement. While there many precision single supply operational amplifiers, such OP213, OP291, OP284, some good single-supply instrumentation amplifiers, highest performance instrumentation amplifiers still specified dual-supply operation.
3.37
AMPLIFIERS SIGNAL CONDITIONING
THREE IN-AMP SINGLE SUPPLY RESTRICTIONS
VSIG VOH=4.9V VOL=0.1V VOH=4.9V VOL=0.1V VREF 2.5V GVSIG VOUT VOH=4.9V VOL=0.1V VOUT= GVSIG VREF GVSIG
VSIG
Figure 3.34
achieve both high precision single-supply operation takes advantage fact that several popular sensors (e.g. strain gauges) provide output signal centered around (approximate) mid-point supply voltage reference voltage), where inputs signal conditioning amplifier need operate near "ground" positive supply voltage. Under these conditions, dual-supply instrumentation amplifier referenced supply mid-point followed "rail-to-rail" operational amplifier gain stage provides very high precision. Figure 3.35 illustrates such high-performance instrumentation amplifier operating single, supply. This circuit uses AD620 low-cost precision instrumentation amplifier input stage, AD822 JFET-input dual rail-to-rail output operational amplifier output stage. this circuit, form voltage divider which splits supply voltage half +2.5V, with fine adjustment provided trimming potentiometer, This voltage applied input AD822 which buffers provides lowimpedance source needed drive AD620's reference pin. AD620's Reference input resistance input signal current 200µA. other half AD822 connected gain-of-3 inverter, that output ±2.5V, "rail-to-rail," with only ±0.83V required AD620. This output voltage 3.38
AMPLIFIERS SIGNAL CONDITIONING level AD620 well within AD620's capability, thus ensuring high linearity "dual-supply" front end. Note that final output voltage must measured with respect +2.5V reference, GND.
PRECISION SINGLE-SUPPLY COMPOSITE IN-AMP WITH RAIL-TO-RAIL OUTPUT
+2.5V
VSIG
10µF
0.22µF 0.1µF 10Hz NOISE FILTER
VSIG
AD620
24.9k
75.0k VOUT 10mV 4.98V
AD822 49.9k
VREF +2.5V
Figure 3.35 general gain expression this composite instrumentation amplifier product AD620 inverting amplifier gains: 49.4 GAIN
this example, overall gain realized with 21.5k (closest standard value). table (Figure 3.36) summarizes various RG/gain values performance. this application, allowable input voltage either input AD620 must between +3.5V order maintain linearity. example, overall circuit gain common mode input voltage range spans 2.25V 3.25V, allowing room ±0.25V full-scale differential input voltage required drive output ±2.5V about VREF.
3.39
AMPLIFIERS SIGNAL CONDITIONING
inverting configuration chosen output buffer facilitate system output offset voltage adjustment summing currents into stage buffer's feedback summing node. These offset currents provided external DAC, from resistor connected reference voltage. AD822 rail-to-rail output stage exhibits very clean transient response (not shown) small-signal bandwidth over 100kHz gain configurations 300. Note that excellent linearity maintained over 0.1V 4.9V VOUT. reduce effects unwanted noise pickup, capacitor recommended across A2's feedback resistance limit circuit bandwidth frequencies interest.
PERFORMANCE SUMMARY SINGLE-SUPPLY AD620/AD822 COMPOSITE IN-AMP
CIRCUIT GAIN 1000
VOS, (µV) 1000
VOS, (µV/°C) 1000
NONLINEARITY BANDWIDTH (kHz)** (ppm)
21.5k 5.49k 1.53k
Nonlinearity Measured Over Output Range: 0.1V VOUT 4.90V Without 10Hz Noise Filter
Figure 3.36
cases where zero-volt inputs required, AD623 single supply in-amp configuration shown Figure 3.37 offers attractive solution. emitter follower level shifters, Q1/Q2, allow input signal 150mV below negative supply within 1.5V positive supply. AD623 fully specified single power supplies between +12V dual supplies between ±2.5V (see Figure 3.38). AD623 data sheet (available http://www.analog.com) contains excellent discussion allowable input/output voltage ranges function gain power supply voltages.
3.40
AMPLIFIERS SIGNAL CONDITIONING
AD623 SINGLE-SUPPLY IN-AMP ARCHITECTURE
VREF VOUT
Figure 3.37
AD623 IN-AMP SPECIFICATIONS
Wide Supply Range: Input Voltage Range: 0.15V 1.5V 575µA Maximum Supply Current Gain Range: 1000 100µV Maximum Input Offset Voltage (AD623B) 1µV/°C Maximum Offset Voltage (AD623B) 50ppm Gain Nonlinearity 105dB 60Hz, Source Imbalance, 0.1Hz 10Hz Input Voltage Noise Figure 3.38
3.41
AMPLIFIERS SIGNAL CONDITIONING
Instrumentation Amplifier Error Sources
noise specifications instrumentation amplifiers differ slightly from conventional amps, some discussion required order fully understand error sources. gain in-amp usually single resistor. resistor external in-amp, value either calculated from formula chosen from table data sheet, depending desired gain. Absolute value laser wafer trimming allows user program gain accurately with this single resistor. absolute accuracy temperature coefficient this resistor directly affects in-amp gain accuracy drift. Since external resistor will never exactly match internal thin film resistor tempcos, (<25ppm/°C) metal film resistor should chosen, preferably with 0.1% better accuracy. Often specified having gain range 1000, 10,000, many in-amps will work higher gains, manufacturer will guarantee specific level performance these high gains. practice, gain-setting resistor becomes smaller, errors resistance metal runs bond wires become significant. These errors, along with increase noise drift, make higher single-stage gains impractical. addition, input offset voltages become quite sizable when reflected output high gains. instance, 0.5mV input offset voltage becomes output gain 10,000. high gains, best practice instrumentation amplifier preamplifier then post amplifier further amplification. pin-programmable gain in-amp such AD621, gain setting resistors internal, well matched, gain accuracy gain drift specifications include their effects. AD621 otherwise generally similar externally gain-programmed AD620. gain error specification maximum deviation from gain equation. Monolithic in-amps such AD624C have very factory trimmed gain errors, with maximum error 0.02% 0.25% being typical this high quality in-amp. Notice that gain error increases with increasing gain. Although externally connected gain networks allow user gain exactly, temperature coefficients external resistors temperature differences between individual resistors within network contribute overall gain error. data eventually digitized presented digital processor, possible correct gain errors measuring known reference voltage then multiplying constant. Nonlinearity defined maximum deviation from straight line plot output versus input. straight line drawn between end-points actual transfer function. Gain nonlinearity high quality in-amp usually 0.01% (100ppm) less, relatively insensitive gain over recommended gain range.
3.42
AMPLIFIERS SIGNAL CONDITIONING total input offset voltage in-amp consists components (see Figure 3.39). Input offset voltage, VOSI, that component input offset which reflected output in-amp gain Output offset voltage, VOSO, independent gain. gains, output offset voltage dominant, while high gains input offset dominates. output offset voltage drift normally specified drift (where input effects insignificant), while input offset voltage drift given drift specification high gain (where output offset effects negligible). total output offset error, referred input (RTI), equal VOSI VOSO/G. In-amp data sheets specify VOSI VOSO separately give total input offset voltage different values gain.
IN-AMP OFFSET VOLTAGE MODEL
RS/2 VOSI
VSIG
VOSO IN-AMP GAIN
VOUT
VSIG RS/2
VREF VOSO VOSI IBRS IOS(RS
OFFSET (RTI)
OFFSET (RTO) VOSO VOSI IBRS IOS(RS
Figure 3.39 Input bias currents also produce offset errors in-amp circuits (see Figure 3.39). source resistance, unbalanced amount, (often case bridge circuits), then there additional input offset voltage error bias current, equal IBRS (assuming that IB). This error reflected output, scaled gain input offset current, IOS, creates input offset voltage error across source resistance, RS+RS, equal IOS( RS+RS), which also reflected output gain, In-amp common mode error function both gain frequency. Analog Devices specifies in-amp source impedance unbalance frequency 60Hz. common mode error obtained dividing common mode voltage, VCM, common mode rejection ratio, CMRR. Power supply rejection (PSR) also function gain frequency. in-amps, customary specify sensitivity each power supply separately. that error sources have been accounted for, worst case error budget calculated reflecting sources in-amp input (Figure 3.40). 3.43
AMPLIFIERS SIGNAL CONDITIONING
INSTRUMENTATION AMPLIFIER AMPLIFIER ERRORS REFERRED INPUT (RTI)
ERROR SOURCE Gain Accuracy (ppm) Gain Nonlinearity (ppm) Input Offset Voltage, VOSI Output Offset Voltage, VOSO Input Bias Current, Flowing Input Offset Current, IOS, Flowing Common Mode Input Voltage, Power Supply Variation, VALUE Gain Accuracy Input Gain Nonlinearity Input VOSI VOSO IBRS IOS(RS CMRR PSRR
Figure 3.40
Instrumentation Amplifier Noise Sources
Since in-amps primarily used amplify small precision signals, important understand effects associated noise sources. in-amp noise model shown Figure 3.41. There sources input voltage noise. first represented noise source, VNI, series with input, conventional circuit. This noise reflected output in-amp gain, second noise source output noise, VNO, represented noise voltage series with in-amp output. output noise, shown here referred VOUT, referred input dividing gain, There noise sources associated with input noise currents IN-. Even though usually equal (IN+ IN), they uncorrelated, therefore, noise they each create must summed rootsum-squares (RSS) fashion. flows through half other half. This generates noise voltages, each having amplitude, INRS/2. Each these noise sources reflected output in-amp gain, total output noise calculated combining four noise sources manner: NOISE RTO) VNO2
3.44
AMPLIFIERS SIGNAL CONDITIONING NOISE RTO) VNO2 VNI2
total noise, referred input (RTI) simply above expression divided in-amp gain, VNO2 VNI2
NOISE
IN-AMP NOISE MODEL
RS/2 IN-AMP GAIN RS/2 NOISE (RTI) NOISE (RTO) VNO2 VNI2 IN2RS2 IN2RS2 VREF VOUT
VSIG
VSIG
VNO2 VNI2
1.57 IN-AMP Bandwidth Gain
Figure 3.41 In-amp data sheets often present total voltage noise function gain. This noise spectral density includes both input (VNI) output (VNO) noise contributions. input current noise spectral density specified separately. case amps, total noise must integrated over in-amp closedloop bandwidth compute value. bandwidth determined from data sheet curves which show frequency response function gain.
In-Amp Bridge Amplifier Error Budget Analysis
important understand in-amp error sources typical application. Figure 3.42 shows load cell which fullscale output 100mV when excited with source. AD620 configured gain using external gain-setting resistor. table shows each error source contributes 3.45
AMPLIFIERS SIGNAL CONDITIONING total unadjusted error 2145ppm. gain, offset, errors removed with system calibration. remaining errors gain nonlinearity 0.1Hz 10Hz noise cannot removed with calibration limit system resolution 42.8ppm (approximately 14-bit accuracy).
AD620B BRIDGE AMPLIFIER ERROR BUDGET
+10V AD620B 350, 100mV LOAD CELL AD620B SPECS +25°C, ±15V VOSI VOSO/G 55µV 0.5nA Gain Error 0.15% Gain Nonlinearity 40ppm 0.1Hz 10Hz Noise 280nVp-p 120dB 60Hz MAXIMUM ERROR CONTRIBUTION, +25°C FULLSCALE: 100mV, VOUT Gain Error Gain Nonlinearity Error 0.1Hz 10Hz Noise Total Unadjusted Error Resolution Error 55µV 100mV 0.5nA 100mV 0.15% 40ppm 550ppm 1.8ppm 1500ppm 40ppm
120dB 1ppm 100mV 280nV 100mV Bits Accurate Bits Accurate
50ppm 2.8ppm
2145ppm
42.8ppm
Figure 3.42
In-Amp Performance Tables
Figure 3.43 shows selection precision in-amps designed primarily operation dual supplies. should noted that AD620 capable single supply operation (see Figure 3.35), neither input output capable rail-torail swings. Instrumentation amplifiers specifically designed single supply operation shown Figure 3.44. should noted that although specifications figure given single supply, amplifiers also capable dual supply operation specified both dual single supply operation their data sheets. addition, AD623 AD627 will operate single supply. AD626 true in-amp differential amplifier with thin-film input attenuator which allows common mode voltage exceed supply voltages. This device designed primarily high low-side current-sensing applications. will also operate single supply.
3.46
AMPLIFIERS SIGNAL CONDITIONING
PRECISION IN-AMPS: DATA ±15V, 1000
Gain Gain Accuracy Nonlinearity 100ppm 0.5% 0.5% 0.05% 0.5% 0.25% 0.02% 0.6% 0.5% 40ppm 10ppm 40ppm 50ppm 50ppm 50ppm 60ppm 50µV 50µV 50µV 125µV 25µV 25µV 50µV 100µV
0.5µV/°C 0.6µV/°C 1.6µV/°C 1µV/°C 0.25µV/°C 0.25µV/°C 0.3µV/°C 2µV/°C
120dB 120dB 100dB 103dB 130dB 125dB 125dB 115dB
0.1Hz 10Hz Noise 0.3µV 0.28µV 0.28µV 0.3µV 0.2µV 0.2µV 0.12µV 0.4µV
AD524C AD620B AD621B1 AD622 AD624C2 AD625C AMP01A AMP02E
Programmable Resistor Programmable
Figure 3.43
SINGLE SUPPLY IN-AMPS: DATA +5V, 1000
Gain Gain Accuracy Nonlinearity 50ppm 100µV AD623B 0.5% AD627B AMP04E 0.35% 0.4% 10ppm 250ppm 200ppm 75µV 150µV 2.5mV 1µV/°C 1µV/°C 3µV/°C 6µV/°C
105dB 85dB 90dB 80dB
0.1Hz 10Hz Supply Noise Current 1.5µV 1.5µV 0.7µV 575µA 85µA 290µA 700µA
AD626B1 0.6%
Programmable Resistor Programmable
Differential Amplifier,
Figure 3.44
3.47
AMPLIFIERS SIGNAL CONDITIONING
In-Amp Input Overvoltage Protection
interface amplifiers data acquisition systems, instrumentation amplifiers often subjected input overloads, i.e., voltage levels excess full scale selected gain range. manufacturer's "absolute maximum" input ratings device should closely observed. with amps, many in-amps have absolute maximum input voltage specifications equal ±VS. External series resistors (for current limiting) Schottky diode clamps used prevent overload, necessary. Some instrumentation amplifiers have built-in overload protection circuits form series resistors (thin film) series-protection FETs. In-amps such AMP-02 AD524 utilize series-protection FETs, because they impedance during normal operation, high impedance during fault conditions. additional Transient Voltage Suppresser (TVS) required across input pins limit maximum differential input voltage. This especially applicable three in-amps operating high gain with values more detailed discussion input voltage EMI/RFI protection found Section this book.
INSTRUMENTATION AMPLIFIER INPUT OVERVOLTAGE CONSIDERATIONS
RLIMIT INPUTS RLIMIT IN-AMP OUTPUT
Always Observe Absolute Maximum Data Sheet Specs! Schottky Diode Clamps Supply Rails Will Limit Input Approximately ±0.3V, TVSs Limit Differential Voltage External Resistors Internal Thin-Film Resistors) Limit Input Current, will Increase Noise Some In-Amps Have Series-Protection Input FETs Lower Noise Higher Input Over-Voltages ±60V, Depending Device) Figure 3.45
3.48
AMPLIFIERS SIGNAL CONDITIONING
CHOPPER STABILIZED AMPLIFIERS
lowest offset drift performance, chopper-stabilized amplifiers only solution. best bipolar amplifiers offer offset voltages 10µV drift. Offset voltages less than with practically measurabl
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