Quasi-Square Wave Resonant converters, often noted converters, offer e

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AND8129/D Power Supply Operating Quasi-Square Wave Resonant Mode
Quasi-Square Wave Resonant converters, often noted converters, offer elegant means make Flyback supplies look more friendly Electro-Magnetic Interference (EMI) point view. delaying switching event until drain-source voltage decayed minimum, switching losses reduced rising slopes lose their stiffness. Designers immediate benefit from this configuration since MOSFET runs cooler input filter becomes easier implement. Designing Switch-Mode Power Supplies (SMPS) requires some attention area dedicated experts only. will discover through following lines Semiconductor solutions help quickly turn your quasi-resonant project into working device.
What Quasi-Resonance?
Ctot
1.69U 1.73U 1.77U 1.81U 1.85U
Figure Truly Resonating Signal Quasi-Resonant Flyback Converter
term quasi-resonance normally related association real hard-switching converter resonant tank. While operation terms control similar that standard controller, additional network added shape variables around MOSFET: current voltage. Depending operating mode, becomes possible either switch zero current (ZCS) zero voltage (ZVS). Compared conventional converter, operation offers less switching losses current circulating through MOSFET increases forces higher conduction losses. However, main advantage favor quasi-resonance reduced spectrum content either conducted radiated. True quasi-resonance means that voltage present switch looks like sinusoidal arch. Figure shows such signal could look.
main problem with this technique lies very high voltage generated switch opening. Most time, these resonant offline designs require around BVdss MOSFETs whose price clearly incompatible with high volume markets. result, designers orientate their choice toward another compromise called quasi-square wave resonant power supplies.
Quasi-Square Wave Resonant Converters
saw, true resonant operation hampers MOSFET selection imposing high voltage switch opening. closely look standard hard-switching waveform (Figure that there exists time where drain voltage gets minimum. This occurs just after core reset.
Semiconductor Components Industries, LLC, 2003
October, 2003 Rev.
Publication Order Number: AND8129/D
AND8129/D
Lleak Ctot
Drain Voltage
Core Reset
Minimum
Ctot
Drain Current
Figure Hard-Switching Waveforms Discontinuous Conduction Mode (DCM)
From Figure possible imagine controller that turns MOSFET until current grows-up setpoint turn then waits until core reset detected (usually auxiliary winding) reactivate this transistor. result, controller does include standalone clock only detects presence events conditioned load/line conditions: this so-called free-running operation. Converters based this technique often designated Self-Oscillating Power Supplies (SOPS), valley switching converters, etc. Oscillations origins seen from Figure arrangement where networks appear. Depending event, different configurations play: switch closing, primary current crosses primary inductance also leakage inductance, Lleak. When turn-on time expires, energy stored transferred secondary side
transformer coupling flux. However, leakage inductance, which models coupling between both transformer sides, reverses voltage imposes quickly rising drain voltage. slope this current
(eq. where Ctot lumps capacitances Ctot
surrounding drain node: MOSFET capacitors, primary transformer parasitics also those reflected from secondary side etc. result, Lleak Ctot form resonating network natural frequency
(eq. maximum drain Lleak Ctot
voltage then computed using characteristic impedance this network.
(Vout Leak (eq. Ctot
Vout Vout
Vout
Lleak
Ctot
tvalley
Multiple valleys.
1.005M
1.015M
1.025M
1.035M
1.045M
Figure Typical Flyback Arrangement Unveils Different Resonating Networks
Figure Typical Flyback Ringing Waveform Occurring Switch Opening
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AND8129/D When transformer core resets, primary
secondary currents drop zero: secondary diode stops conduction reflected voltage primary naturally dies out. From this implies that terms after collapse zero tends toward Vin. However, transition would brutal lack resonating network, this time made primary inductance, nearly same Ctot before. imagine, sinusoidal ringing takes place, damped presence ohmic losses resistance primary winding modeled Rp). drain-source shape rings below formula details:
Vds(t) (Vout cos(2 fprim
(eq.
input voltage, diode's forward drop Ns:Np turn ratio. from Figure that drain seat various voltage drops when going down ringing wave. These drops called "valleys". manage switch MOSFET right middle these valleys, ensure minimum turn-on losses, particularly those related capacitive dissipation:
Pavgcap Ctot Vds2 (eq. Thus, quasi-
with:
(eq. (the damping factor), fprim
square wave operation valley switching, will imply reactivation switch when minimum. various figures portray, this occurs some time further transformer core reset. implementing this method, build converter which naturally exhibits variable frequency operation since reset time depends upon input/output operating conditions. Figure shows typical shot quasi-square wave converter.
(eq. Ctot
(natural ringing frequency),
Ipeak (Vout
Valley
Figure Typical Drain-Source Shot Quasi-Square Wave Converter
Figure Primary Inductance Current Made Different Slopes (here restart occurs second one)
see, total period made different events, where core first magnetized (Ton), then fully reset (Toff) finally time (Tw) delay inserted reach lowest value drain. look frequency moves respect input/output conditions.
Evaluating Free-Running Switching Frequency
event, which fourth natural ringing frequency given equation will compute derivative equation null find minimum:
d(Vin cos(2 fprim (eq.
Which gives result
fprim
free-running frequency evaluated looking Figure where primary current (circulating primary inductance) depicted. From definition various slopes, express first events, Toff quite easily:
toff VinDC
fprim
(eq.
(eq.
(Vout
However, this result very practical because inherent complexity. observe equation that minimum reached when term cos(2 fprim equals Otherwise stated, solve which cosine null full product equals This gives:
fprim (eq.
(eq.
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AND8129/D
However, this result valid only damping coefficient, that say, Experience shows that good enough vast majority cases. result, final switching period computed summing these sequences introducing input power expression:
Toff Toff VinDC
(eq. (eq. (eq.
(Vout
(eq.
Pout from Pout
converter efficiency Pout output power Vout respectively output voltage rectifier drop Iout primary inductance Stating that Pout
Now, plugging (eq. gives:
(eq. 16a) Pout VinDC Vreflect [Vout Vreflect
with: Vreflect
Vreflect2 Vreflect Vin2 Vin2 Vreflect2 Vreflect) Pout
(Vout)Vf))Vin (Vin (Vout)Vf))))
(eq.
From equation then compute switching frequency using calculated peak current:
Pout
(eq.
(eq.
However, equation very practical since involves what actually looking for. certainly used discover operating peak current from known inductance capacitor values neglecting offers simpler formula that used first frequency iteration (e.g. feed Spice simulator instance):
7*104 6*104
Entering equation into spreadsheet plotting versus various parameters (Vout, Iout etc.), gives idea about high frequency variability system. Figure Figure respectively plot function input voltage output current given application.
2*105
1.5*105 5*104 f(VinDC) 4*104 3*104 5*104 2*104 1*104 f(Po) VinDC 1*105
OUTPUT POWER
Figure Frequency Variations SMPS Operated from Universal Mains
Figure Frequency Dependency with Load Given Input Voltage (100
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AND8129/D
Ip(VinDC)
VinDC
Figure Peak Current Variations Output Power with Different Line Voltages Quiet Signature
Manipulating sinusoidal close-to) variables always offer narrower spectrum content compared hard-switching systems. Figures depict
conducted signature systems operated same point implementing different switching techniques.
Figure soft-switching approach reduces energy content above
Figure while hard-switching system generates noise this portion Detecting Core Reset Event
Since MOSFET reactivated lowest drain level, classical Coss capacitor discharge switch closing non-existing very narrow peak current gone (also this peak often confusing current-sense comparator when really energetic, even sometimes despite presence circuitry). result, Quasi-square wave converters recommended where Switch-Mode Power Supply (SMPS) needs operate close Radio-Frequency section, e.g. Boxes, sets, etc.
Core reset detection usually done dedicated auxiliary winding whose voltage image directly linked transformer flux Vaux
(eq.
Depending controller device, polarity observed signal must detection circuitry. Semiconductor NCP1205, this polarity should Forward type, that say, when MOSFET opens, auxiliary voltage (actually Flyback level) dips below ground stays there, safely clamped -0.7 until core reset occurs. Figure gives example demagnetization signal given auxiliary winding wired both types.
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AND8129/D
20.0 10.0 10.0 20.0
N.Vin
Flyback operation
20.0 10.0 10.0 20.0
N.Vin
Forward operation 65mV
Leakage contribution Watch possible start!
Figure Core Reset Detection Signal Coming From Either Forward Flyback Winding
Operating auxiliary winding Forward offers various advantages variable level introduce overpower compensation. Also controller always operating (supplied) whatever secondary output conditions. Figure shows possible that. Please note presence small filter necessary introduce time delay after core resets (and thus activate MOSFET right minimum valley wave) filter leakage contribution that could adversely restart controller higher switching frequency (see Figure evidence). Overpower compensation there avoid larger over current trip point high-line compared low-line conditions. instance, suppose that maximum peak current line (100 would pass that your maximum peak current (given sensing element internal clamping setpoint, usually
fixed means that overcurrent condition exists soon output load slightly increases, perhaps which what defined with maximum peak Now, converter high line, e.g. VDC, peak current will decrease, shown Figures thru down result, still have dynamic before hitting maximum current trip point. SMPS thus theoretically deliver before actually trips. overcome this problem, wire resistor between current sense since, forward polarity,
Naux VinDC (eq. which direct image
mains. result, moves with high-voltage rail offsets current sense reading, offering natural, power, input feedforward.
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AND8129/D
Demag Rlimit Line Operation Overpower
Vout
High Line Operation
Figure Quasi-resonant applications impose different operating peak current depending input line.
Figure Wiring auxiliary forward mode offers ability build inexpensive overpower compensation since aux. moves with input voltage.
Care should taken however, inject much over power level into otherwise affect operation. some applications, difficult cope with variable auxiliary level Flyback option better.
adding second diode, becomes easy wire auxiliary winding Flyback mode, still offering core reset signal Forward polarity. Figure offers example, where demagnetization signal undergoes high-frequency filtering network.
NCP1205 Quasi-Resonant Controller
Vout
Demag
Rlimit
Rsense
Figure Wiring Winding Flyback Mode also possible with NCP1205
This NCP1205 available DIP8, DIP14 SO-16, offers many features that make right candidate quasi-resonant applications: Full Quasi-Square Wave Resonant Operation: detecting transformer core demagnetization initiate cycle, NCP1205 ensures drain-source valley switching operation. Furthermore, comprehensive logic circuitry, device jumps between valleys built-in starts decrease switching frequency. result, Electromagnetic- Interference (EMI) reduced turn-on losses virtually null. Voltage-Controlled Oscillator: internal takes over soon free-running frequency hits maximum user adjustable value. output power demand further diminishes, switching frequency naturally reduced ensure better efficiency light loads.
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AND8129/D Standby Power: SMPS naturally exhibits
good efficiency nominal load, they begin less efficient when output power demand vanishes. smoothly reducing number switching cycles second, NCP1205 drastically reduces power wasted during light load conditions. no-load conditions, NCP1205 allows total standby power easily reach exceed next International Energy Agency (IEA) recommendations. Short-Circuit Protection: permanently monitoring feedback line activity, able detect presence short-circuit, immediately reducing output power total system protection. Once short disappeared, controller resumes goes back normal operation. given applications, easily disconnect this protective feature. This short-circuit detection independent from auxiliary level, hence lose coupling between auxiliary power windings problem. Overvoltage Protection: continuously checking rail, NCP1205 safely into latch-off phase when operating voltage exceeds Forward winding applications, this options lets also protect design against transient mains over voltages. application where adjustment necessary, DIP14 versions pins dedicated comparator input select protection level your choice. Large Supply Range: Battery charger applications require that controller still control output current when output voltage close zero (e.g. discharged battery). This called Constant-Current/ Constant-V oltage (CC-CV) operation. allow controller self-supply when output voltage disappears, needs wire auxiliary winding Forward mode. However, most today's primary side controllers have difficulty cope with Forward auxiliary winding operated universal mains because large voltage dynamics implies. Fortunately, authorizing through operation, NCP1205 eases designer task self-supply side. Output Ripple Standby: Some loads sensitive ripple present output. This case Li-Ion batteries where clean voltage required ensure longest service. Standard hysteretic controllers produce un-acceptable output ripple. smoothly reducing operating frequency, NCP1205 generates lower ripple when entering standby mode. Acoustic Noise While Operating: Instead reducing switching frequency high peak currents, NCP1205 waits until peak current demand falls below fixed 1/3rd peak maximum limit. result, frequency reduction takes place without having singing transformer. thus select cheap magnetic components free noise problems.
External MOSFET Connection: leaving
external MOSFET external select avalanche proof devices which, certain cases (e.g. output powers), work without active clamping network. Also, controlling MOSFET gate signal flow, have option slow down device commutation, therefore reducing amount ElectroMagnetic Interference (EMI). SPICE Model: dedicated model that lets transient cycle-by-cycle simulations available verify your theoretical design. Ready-to-use templates downloaded OrCAD's PSpice INTUSOFT's from Semiconductor site, NCP1205 related section. Complete details regarding implementation NCP1205 given application note AND8043.
Power Supply Design Using Quasi-Resonant Approach
Pout nominal Vout 16.8 Universal input voltage 90-265 (including losses ripple) Short-circuit protection Standby power less than no-load design quasi-resonant converter featuring standby power requires understanding several parameters before calculating anything: need ensure true Zero Voltage Switching (ZVS) operation over large operating input range? ensure high switching frequency maximum power mains, while minimizes magnetics, frequency foldback will eventually take place higher input voltages. what peak current level authorize frequency foldback avoid acoustical noise transformer?
Answer
Yes, because will shape drain-source voltage (especially switch opening) smooth possible soften signature wire large resonating capacitor whose losses should minimized (see will save costly clamping network. result, will reflect much can, taking into account BVdss MOSFET higher secondary rectifier losses (peak secondary currents up). have selected turn ratio 16.6 gives reflected voltage drain stress high line without leakage, thus: (265 1.414) which gives room leakage effects. Please remember that best reflect maximum voltage from secondary side, best case being Vreflect (Vout Vin. that
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AND8129/D
latest case, would probably need pick BVdss MOSFET.
Answer
case, prefer avoid foldback nominal load over whole input voltage. Frequency foldback starts discrete jumps between valleys create some noise. accept increase frequency high line before folding frequency back (with connected clamp Fmin kHz), then accept lower switching frequency lines, avoid entering audible frequencies.
Answer
answer really depends upon transformer structure have used, e.g. type core, bobbin, etc. best setup test structure where impose peak current your transformer prototype low, audible, frequency (e.g. around kHz). Figure offers possibility that power MOSFET.
generate burst pulses drive MOSFET. discontinuity associated with burst sequence more favorable trigger mechanical resonances compared evenly spaced pulses. Suppose that peak current offering best value with your transformer. Since 1205 folds back maximum primary current, will select maximum peak current V/1.65 Rsense Iterations using dedicated Excel spreadsheet will therefore help select right primary inductance turns-ratio reach good performance standby without making noise. Let's follow below design steps build Switch-Mode Power Supply: opinion, very first element dimension primary secondary turn ratio. effect, will condition, among other parameters, drain-source stress MOSFET opening Peak Inverse Voltage (PIV) secondary rectifier switch closing area where supply operates Zero Voltage Switching (ZVS). have seen before, select MOSFET, select turn ratio (including safety margin):
VinDC (Vout V-10%
(eq.
PULSE
Current Reading
Np/Ns 19.5. selected 16.6 turn-ratio which will ensure 16.6 (16.8 VDC. Having right turn ratio, calculate primary peak current needed pass power. neglect delay reach valley Vds(t) (see 12), then with simplified current definition:
Pout
(Vout VinDC
VinDC
(Vout
Figure power MOSFET adjustable duty-cycle generator lets select right peak current.
adjusting PULSE source duty-cycle, becomes possible impose given peak current, directly sensed across resistor oscilloscope. freewheel diode could 1N4937/MUR160 equivalent whereas source around with BVdss MOSFET. Start peak currents slowly increase duty-cycle until noise heard. This corresponds very maximum peak current pass while skipping cycles when entering frequency foldback without having singing transformer. best actually
plugging values into gives maximum peak current 0.94 This value will slightly change soon consider other parasitic elements (see AND8089, "Determining Free-Running Frequency Systems"), good starting point. From that value, know that NCP1205 will start folding back frequency into audible range peak current equal maximum value (this NCP1205 designed). know experience (see Figure 16), that shall over avoid having singing transformer. case, 0.94 well within specs. even have place improvement feel need increase parameter variation reasons.
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AND8129/D
inductor defined knowing what frequency range want cover. exemplified Figure switching frequency increases high input voltage (whereas goes low) decreases input voltages (whereas goes up). some cases, desirable keep magnetics small thus operate high frequency line. other way, some designers find that desirable compensate higher losses line, reducing switching losses lower switching rate. will stick this latest option calculate above audible range line maximum output power. Rearranging equation leads solve:
Pout
(Vout)Vf))Vin (Vout)Vf)
(eq.
greater than This number first result will that further iterations needed freeze this number. Since implement true connect large capacitor between drain ground clamp maximum voltage generated leakage inductance. capacitor losses will null (see will start increase high line when lost. believe that even detrimental efficiency, cost improvement brought absence clamping network smoother waveforms (good EMI), really justifies addition this drain-ground capacitor. tweaking equation calculate amount necessary capacitance need between drain ground:
Creso Lleak
-Vin-
obviously most sensitive parameters which influences others. Increasing reflected voltages keep wider operating range price other numbers: switching frequency increases (reset voltage stronger) primary peak current conduction losses improved goes peak demand goes low) secondary peak current conduction losses increase MOSFET undergoes bigger stress switch opening MOSFET turn-on losses really null achieved) Final values will obtained design spreadsheet available download from Semiconductor site which includes parasitic elements (such leakage inductance capacitor) whose formulae described AND8089. After entered desired operating conditions, below numbers were extracted from spreadsheet: Lleak (measured) Np/Ns 16.6 include various tolerances Rshunt parallel Creso nF/1.0 Calculated frequency nominal load minimum input voltage (120 Calculated frequency nominal load maximum input voltage (370
Using SPICE Check Validity Assumptions
(Vout
(eq.
consider leakage inductance around (first estimation) plug values into equation then Creso needs greater than losing imagine that switch restart will occur drain wave V-295 These nominal conditions imply theoretical switching frequency (eq. capacitive losses (eq. Ploss which acceptable. through lines wrote that many parameters changed obtain different converters end. reflected voltage
Despite existence dedicated NCP1205 SPICE model, faster easier simplified free-run approach have idea final results. Figure offers possible represent free-running controller: demagnetization path includes standard flip-flop which latches transition while feedback signal fixes current setpoint. simple arrangement, system simulates really quickly allows immediate assessment what been suggested Excel spreadsheet. feedback loop purposely simplified with Zener diode arrangement, upgrade with TL431 circuitry. will simply take longer simulation time settle. Figures show, difficult make distinction between simulation real measurement demoboard.
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AND8129/D
XFMR-AUX RATIO_POW -0.06 RATIO_AUX 0.06 Iout MBR20100 Idiode Vout Vout
Rprim
Icoil VCoil Lprim
Resr1 Cout1
Rload
Lleak
Free Free
Vdrain Feedback IReso Creso MOC8101 15.6 Vout Rled
Rsense
Figure Simplified Free-Running Controller Eases Simulation Setup Increases Speed
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AND8129/D
V/div
Vsense mV/div ms/div
Figure
Figure
becomes difficult differentiate simulation from real world Figure simulation, Figure measured.
Spice simulation offers another advantage which evaluation component stresses. good models, immediately measure MOSFET conduction losses worse case, current rectifiers, resonating capacitor, etc. Figure portrays these typical waveforms. light this data, select components peripherals accordingly: MOSFET: Depending type, compute power using: Pconduction RDS(ON) 100°C IdRMS2 0.4652 line. case, switching losses close zero ZCS. When main grows bulk capacitor, effect goes away capacitive losses appear Creso. Simulation shows that conduction losses stay below selected MOSFET from small note about MOSFET Spice models: They reflect temperature effects RDS(ON) other parameters (e.g. Vth). result, calculating total power (including switching losses) multiplying averaging Vds(t) Id(t) over period does only make sense junction temperatures 27°C.
Resonating Capacitor: This device shall sustain voltage peak also large current. Simulations show that worse case occurs high line with current have used WIMA FPK1 series with good results. Primary Inductance: line imposes highest stress transformer. following specs passed transformer manufacturer: Ipmax IpRMS Np:Ns 1:0.06 Np:Naux 1:0.06 Secondary Rectifier: conduction losses diode given IdRMS2 IdAVG. case, obtain theoretical total losses 1.21 MBR10100 good choice.
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AND8129/D
1.20 plot1 amperes 800m 400m icoil idiode ireso i(cout1)
-400m Plot2 icoil amperes 1.20 800m 400m
Idrain 465mA
-400m 28.7 21.6 14.6 7.51 434m 1.00 600m 200m
Icoil 571mA
Plot3 idiode amperes
5.02A
Plot4 ireso amperes
ICreso 223mA
-200m -600m 30.0 20.0 10.0 -10.0 2.01m 2.03m 2.05m Time Seconds 2.07m
Plot5 i(cout1) amperes
ICout 5.72A
2.09m
Figure Typical Waveforms Obtained Spice Simulation Quasi-Resonant Converter
Output Capacitor: This component will selected based required output ripple also ability sustain total current. dissipation capacitor dictated Equivalent Series Resistor (ESR) follows following formula: Pcap RESR Iripple2. current will thus primary criterion when selecting right device. also possible wire capacitors parallel split total current between devices.
Final Schematic
Figure shows final schematic implemented NCP1205 demoboard. see, large capacitor placed drain allows avoid costly noisy clamping network. However, layout offers necessary place include experiments needed. auxiliary winding wired offer stable Flyback
voltage diode (D5) placed series with ground generate right Forward polarity (see Figure 15). Because resonating capacitor placed between drain ground, spike occur high line soon effect lost. help circuitry inside NCP1205, additional cleaning network added through R2-C2. Please note that resonating capacitor wired between drain ground between drain source. This avoid negative current flowing inside current sense turn-off (during natural drain-source ringing). feedback loop standard uses TLV431 further lower secondary side standby power. Since minimum operating current there need waste with traditional TL431 series.
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1N4148 Cclamp 2KBP08M Wired Schaffner RN114-08/02 Universal Input Rclamp
1:0.06 (power) 1:0.06 (aux.)
Coilcraft PCV-2-103-05 MBR20100
13.5 Ground
Figure Final Electrical Diagram NCP1205-Based Demoboard
Dclamp Optional Network
1000 mF/25 C15/C16/C17
nF/1 WYMAFKP1 1000
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AND8129/D
NCP1205
Optional Network
SFH6156-2 Type
TLV431
AND8129/D
TLV431 really reduces output power wasted no-load. sometimes, repetition rate switching pulses standby low, that auxiliary goes down reaches NCP1205 UVLOlow, restarting high-voltage current source. avoid this situation, either need increase auxiliary turn ratio (this problem because extended range offers that flexibility) slightly load output resistor. Experience shown that when auxiliary close simple bleeding element connected Vout enough bring Vauxiliary around giving necessary headroom.
Demoboard Performance Typical Waveforms
demoboard available from Semiconductor. corresponds Figure sketch. Below some shots measurements captured board, testifying good characteristics: Efficiency 86.6% Standby power (Pout Efficiency 84.4% Standby power (Pout
Figure Drain-source signals different powers: tP4, note multiple valley jumping.
Figure high line nominal power, drain level kept below MOSFET breakdown.
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AND8129/D
Figure high line, output short circuit does jeopardize MOSFET's life.
Figure output bang-bang test does reveal instability. spikes related filter installed output.
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AND8129/D
Bill Materials resistors unless otherwise noted.
Part Value mF/400 nF/1.0 1000 mF/25 1000 mF/25 1000 mF/25 mF/25 mF/25 BRIDGE MBR20100 STP5NB80 NCP1205P SFH6156-2 TLV431 Transformer Designator Roederstein Coilcraft Schaffner KBU-8J Semiconductor Semiconductor Infineon Semiconductor Coilcraft Pulse-Engineering Z9508-A PF0137 TO-92 Series PCV-2-103-05 RN114-08/0.2 Vertical Vertical Vertical Wima 1500-FKP1 Snap-in Comp. 2222-057-56479 Through Holes Through Holes Through Holes Type Manufacturer Reference Comments
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AND8129/D
Transformer Vendor Details Pulse Engineering Site d'Orgelet Zone industrielle 39270 ORGELET Tel.: (0)3 Fax: (0)3 http://www.pulseeng.com/ Email: vpelletier@pulseeng.com Coilcraft 1102 Silver Lake Road Cary, Illinois 60013 Tel.: (847) 639-6400 Fax: (847) 639-1469 Email: info@coilcraft.com http://www.coilcraft.com
Semiconductor registered trademarks Semiconductor Components Industries, (SCILLC). SCILLC reserves right make changes without further notice products herein. SCILLC makes warranty, representation guarantee regarding suitability products particular purpose, does SCILLC assume liability arising application product circuit, specifically disclaims liability, including without limitation special, consequential incidental damages. "Typical" parameters which provided SCILLC data sheets and/or specifications vary different applications actual performance vary over time. operating parameters, including "Typicals" must validated each customer application customer's technical experts. SCILLC does convey license under patent rights rights others. SCILLC products designed, intended, authorized components systems intended surgical implant into body, other applications intended support sustain life, other application which failure SCILLC product could create situation where personal injury death occur. Should Buyer purchase SCILLC products such unintended unauthorized application, Buyer shall indemnify hold SCILLC officers, employees, subsidiaries, affiliates, distributors harmless against claims, costs, damages, expenses, reasonable attorney fees arising directly indirectly, claim personal injury death associated with such unintended unauthorized use, even such claim alleges that SCILLC negligent regarding design manufacture part. SCILLC Equal Opportunity/Affirmative Action Employer. This literature subject applicable copyright laws resale manner.
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AND8129/D

Quasi-Square Wave Resonant converters, often noted converters, offer e

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