SiP12510/11 SiP12511 SiP12510 TSOT23-6 MBR0540 MBR0530 S-70770 SiP12510DT-T1-E3 - Datasheet Archive

 

SiP12510/11 Vishay Siliconix 1.25-MHz Boost Converter White LED Driver with Internal Power Switch APPLICATIONS · ·

 

New Product SiP12510/11 SiP12510/11 Vishay Siliconix 1.25-MHz Boost Converter White LED Driver with Internal Power Switch APPLICATIONS · · · · · · Portable Phones and Game Devices PDAs and Palm-Top Computers Local Boost Regulator CCD Bias Supplies Digital Cameras TFT-LCD Displays · · · · DSL Modems PCMCIA Cards White LED Backlight OLED Driver DESCRIPTION The SiP12510/11 SiP12510/11 are a 1.25 MHz current-mode boost converter with a feedback reference voltage of 0.1 V which offers small size and high power conversion efficiency. Its input voltage range is from 2.5 V to 6 V, and output voltage can go up to 27.5 V for SiP12511 SiP12511 and 17 V for SiP12510 SiP12510. The internal frequency compensation minimizes number of external components. The integrated 33 V power switch can carry up to 0.55 A. The integrated power switch also features the over current limiting to protect itself. The internal soft-start circuit controls the rate of rise of the output voltage during start-up to prevent overshoot. The logic-level shutdown pin can be used to reduce quiescent current to < 1 µA and, effectively, extend battery life. Thermal shutdown at 165 °C is also included. The low FB voltage of 0.1 V improves the overall circuit efficiency. These features and more, make the SiP12510/11 SiP12510/11 an ideal power solution to white LED, OLED, LCD, and CCD applications operating from a single or dual cell lithium-ion battery. SiP12510/11 SiP12510/11 are available in 6-pin TSOT23-6 TSOT23-6 package and are specified to operate over the industrial temperature range of - 40 °C to 85 °C. FEATURES · Output Voltage Range up to 27.5 V for · · · · · · · · · · · · SiP12511 SiP12511 and 17 V for SiP12510 SiP12510 Current Mode Control with Internal Frequency Compensation 2.5 V to 6 V Input Voltage Range 1.25 MHz Switching Frequency Low Shutdown Current (< 1 µA) Under Voltage Lockout Protection Output Over Voltage Protection Thermal Shutdown Protection (165 °C) 0.55 A Switch Current Limiting High Efficiency up to 90 % Built-in Soft Start Control Minimum External Components TSOT23-6 TSOT23-6 Package RoHS COMPLIANT TYPICAL APPLICATION CIRCUIT MBR0540 MBR0540 VIN VOUT L 10 µH CIN 1 µF SHD COUT 1 µF 6 VIN 4 LX SHD 1 SiP12511 SiP12511 5 FB VOUT 3 GND 2 0.1 V RFB = 5 Figure 1. SiP12511 SiP12511 Typical Application Circuit MBR0530 MBR0530 L VIN VOUT 10 µH CIN 1 µF SHD COUT 1 µF 6 VIN 4 LX SHD 1 SiP12510 SiP12510 5 FB VOUT 3 GND 2 0.1 V RFB = 5 Figure 2. SiP12510 SiP12510 Typical Application Circuit Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 www.vishay.com 1 New Product SiP12510/11 SiP12510/11 Vishay Siliconix ABSOLUTE MAXIMUM RATINGS Parameter Limit Unit Input Voltage, VIN to GND - 0.3 to 12 VOUT, LX Voltage - 0.3 to 33 SHD Voltage - 0.3 to 12 - 0.3 to 12 FB Voltage ESD (Human Body V Model)a 2 Maximum Junction Temperature kV 150 Storage Temperature °C - 55 to + 150 Power Dissipation (TA = 70 °C)b 367 Junction to Ambient Thermal Impedance (RJ) °C/W 125 Maximum Operating Junction Temperature mW 150 c °C Notes: a. The human body model is a 100 pF capacitor discharged through a 1.5 k resistor into each pin. b. Derate 6.67 mW/°C above 70 °C. c. Device mounted with all leads soldered or welded to PC board. Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating/conditions for extended periods may affect device reliability. RECOMMENDED OPERATING RANGE Parameter Input Voltage, VIN to GND 2.5 to 6 Limit SHD 0 to VIN SiP12511 SiP12511 VOUT V SiP12510 SiP12510 VIN to 27.5 LX Unit VIN to 17 V 0 to 28 Operating Temperature Range - 40 to 85 °C SPECIFICATIONS Parameter Input Voltage Switch Current Limit Test Conditions Unless Specified VIN = 5 V, VSHD = 2 V, TA = 25 °C Temp Mina VIN Full 2.5 ILIMIT Full 0.385 Symbol ISW = 100 mA RDS (on) SHD Input High Level VSHDH Full SHD Input Low Level VSHDL VFB Full Feedback Bias Current Feedback Voltage Line Regulation V 0.55 0.735 A 1.25 0.4 V 0.100 0.108 1.5 0.092 60 5 VIN 2.5 V and VOUT = 15 V at 20 mA nA 0.2 IFB VFB/ (VFB x VIN) Unit 6 Full Feedback Voltage Maxa 0.75 Full Switch On Resistance Typb %/V VFB = 0 V (Switching) IQ Switching Frequency 1.3 2 Full 0.3 0.5 Full 1 µA 1.45 MHz FSW Maximum Duty Cycle Full VFB = 1.5 V (Not Switching) VSHD = 0 V Quiescent Current DMAX Switch Leakage ILEAK Thermal Shutdown TSHD www.vishay.com 2 Full 1.25 Full Not Switching, VLX = 5 V 1.05 85 90 % Room 1 Full 10 165 mA µA °C Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 New Product SiP12510/11 SiP12510/11 Vishay Siliconix SPECIFICATIONS Thermal Shutdown Hysteresis THYST Under Voltage Lockout VUVLO UVLO Hysteresis 20 2.00 SiP12510 SiP12510 Full 17 19 21 SiP12511 SiP12511 Full 27.5 30 32 VUVLOHYST Over Voltage Protection 2.48 0.1 VOVLO OVLO Hysteresis 2.24 °C Full V 0.2 VOVLOHYST Notes: a. Limits are guaranteed by testing. b. Typical values are derived from the mean value of a large quantity of samples tested during characterization and represent the most likely expected value of the parameter. PIN CONFIGURATION TSOT23-6 TSOT23-6 Package LX 1 6 VIN GND 2 5 VOUT FB 3 4 SHD TOP VIEW Figure 3. PIN DESCRIPTION Pin Number Name Function LX Drain Pin of the Internal Switch. Connect inductor/diode to LX. Minimize trace area at this pin to keep electromagnetic interference down to a minimum. 2 VIN Analog and power input of the controller IC. A bypass capacitor is required on this pin. 3 SHD 1 Logic Controlled Shutdown Input. SHD = high: Normal operation. SHD = low: Shutdown. 4 FB 5 VOUT Voltage Feedback Pin, the inverting input of the voltage error amplifier. This is internally compared against a voltage of 0.1 V appearing on the voltage error amplifier's non-inverting input. External resistors are connected to this pin to set the regulated output voltage. Output Voltage Pin, Output voltage sense for over voltage protection and the slop compensation. 6 GND Signal and Power Ground, this pin acts as both the analog ground and the power ground for this part. ORDERING INFORMATION Part Number SiP12510DT-T1-E3 SiP12510DT-T1-E3 SiP12511DT-T1-E3 SiP12511DT-T1-E3 Marking M3WXX M4WXX Temperature Range Package - 40 °C to 85 °C TSOT23-6 TSOT23-6 XX = Lot Code W = Work Week code Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 www.vishay.com 3 New Product SiP12510/11 SiP12510/11 Vishay Siliconix FUNCTIONAL BLOCK DIAGRAM VIN VOUT LX Thermal Shutdown Over Voltage Lockout Under Voltage Lockout R FF FB Gm PWM Ramp Generator R Q Driver R 0.1 V Reference S Current Limit Comparator Softstart Current Sense SHD Shutdown Control 1.25 MHz Oscillator GND Figure 4. Internal Block Diagram DETAILED OPERATION DESCRIPTION The SiP12510/11 SiP12510/11 is a current mode, internally compensated, step-up switching converter that operates at a fixed frequency of 1.25 MHz. The current mode topology allows for fast transient response over a wide input range and provides a real-time, cycle-by-cycle current limiting function. The operation of the converter can be described through the interaction of two separate internal loops: the current sense loop and voltage sense loop. Within the current sense loop, the switch MOSFET current is monitored by sensing the voltage across an internal current sense resistor, which is fed to the inputs of both the current limit amplifier and the pulse width modulation (PWM) comparator. At the beginning of each switch cycle, the oscillator sets the S-R latch thereby turning on the MOSFET. As current through the switch increases, so does the voltage drop across the sense resistor. This voltage is summed with the ramp coming from the ramp generator and applied to the input of the PWM comparator. When this ramping voltage exceeds the output of the error amplifier, the latch changes state and turns off the MOSFET. The slope of the ramp generator is proportional to voltages on the VIN and VOUT pins, therefore, any sudden changes in input or output voltage can be corrected and accommodated for on a cycle-by-cycle basis. If the MOSFET current surpasses the current limit threshold, the current limit comparator will unconditionally turn off the internal power switch. At the beginning of the next oscillator cycle, the switch is allowed to turn on again. The voltage feedback loop works by monitoring the LED www.vishay.com 4 drive current through a resistor divider on FB and comparing that voltage with an internal 0.1 V reference voltage (Vref). If the LED current falls below the set current, the voltage on the feedback pin will drop slightly below Vref causing the output of error amplifier to increase. This will keep the PWM comparator's output high for a greater portion of an oscillator cycle, thus ensuring that the MOSFET will stay on longer. This means that the duty cycle of the LED driver converter will increase. The output voltage of LED driver converter will increase. This, in turn, will allow more current to be delivered to the load. Following similar logic, should the LED current become higher than the set current, FB voltage will increase above Vref, the converter will decrease its duty cycle, which will lessen the energy delivered to the load at each cycle, and thereby, reduce LED current and maintain desired brightness. In essence, by modifying the on time of the switch, the PWM comparator continually sets the correct maximum current through the MOSFET to regulate the LED current to a desired value. SIP12510/11 SIP12510/11 FUNCTIONAL DESCRIPTION SiP12510/11 SiP12510/11 is the combination of the high voltage low-side MOSFET and the current mode PWM controller. The current mode PWM controller consists of error amplifier, 0.1 V reference voltage, 1.25 MHz oscillator, ramp generator, current sense amplifier, PWM, and current limit comparator, thermal protection, over voltage protection and MOSFET driver. Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 New Product SiP12510/11 SiP12510/11 Vishay Siliconix OSCILLATOR The typical oscillator frequency is internally set to 1.25 MHz. The output of the oscillator is not only used to drive the input of the built-in MOSFET driver. It is also set the operating frequency of the ramp generator for the slop compensation. 0.1 V REFERENCE VOLTAGE The 0.1 V reference voltage is connected on the noninverting input of the error amplifier for output voltage regulation. The typical application of the SiP12510/11 SiP12510/11 boost converter is to driver the white LED. The output voltage of the boost converter is set by the forward drop voltage of the LED in series in the feedback voltage divider. The LED load is in series with the resistor of the voltage divider. The lower the reference is, the higher the efficiency of the converter circuit. Based on the efficiency measurement, the efficiency of the converter can achieve 90 % (see efficiency curve in the typical waveform section). So the efficiency is benefited from 0.1 V low reference voltage. CURRENT LIMIT COMPARATOR AND CURRENT SENSE AMPLIFIER The current limit comparator and the current sense amplifier are design to protect the built-in MOSFET and the converter from over current operation condition. The current sense amplifier not only monitors the switch current for cycle-bycycle current mode operation. It also senses the switch current for cycle-by-cycle over current protection for the built-in switch. The typical maximum over current protection threshold is set to 0.55 A. Once the maximum current exceeds 0.55 A, the current limit comparator will shut down the gate drive signal for the built-in MOSFET to protect the built-in MOSFET. OVER VOLTAGE PROTECTION SiP12510/11 SiP12510/11 have a built-In output voltage protection to shut down the controller. The minimum over voltage protection threshold is 27.5 V for SiP12511 SiP12511 and 17 V for SiP12510 SiP12510. The protection will completely shut down the controller and restart if the over voltage fault occur. The maximum voltage rating for the built-in power MOSFET is 33 V for both SiP12510/11 SiP12510/11. Any potential over voltage on the output of the converter will not be able to damage built-in MOSFET. In the event of an output open circuit (e.g. when the LEDs are either disconnected form the output or an LED fails), the feedback voltage will become zero causing the SiP12510/11 SiP12510/11 to go to maximum duty cycle. This would generally result in a high output voltage and, possibly, cause the voltage on the LX pin to exceed it's absolute maximum rating and damage the part. However, the SiP12510/11 SiP12510/11 have a built-in over-voltage protection circuitry that will clamp the output to 19 V typical for SiP12510 SiP12510 and 30 V typical for SiP12511 SiP12511. SiP12510/ SiP12510/ 11 guarantee safe operation under open-circuit conditions. THERMAL PROTECTION The thermal protection circuit senses the die temperature. The temperature threshold is set to 165 °C typical with 20 °C hysteresis. The built-In MOSFET will be disabled when the temperature exceeds 165 °C remain disabled until the die temperature drop below 145 °C to re-enable. START UP AND SOFT-START When voltage is applied to the VIN pin, the under voltage lockout (UVLO) circuit prevents the controller's output switch and oscillator circuit from turning on until the voltage on the VIN pin exceeds 2.24 V. Provided the VIN pin is above this threshold, when SHD pin is raised high, soft-start is initiated. Soft-start is achieved by slowly ramping up the internal reference. For a certain period of soft-start time (about 0.7 mS), the value of over-current protection threshold is being changed twice: it is about 40 % of its steady state value during the first phase of this period of soft-start time, and about 66 % of its steady state value during the second phase of soft-start period of time. The heavy load is applied on the output of SiP12511 SiP12511 WLED driver converter to show the current limiting phase in soft-start period in Figure 5. Once the softstart time has elapsed, SiP12510/11 SiP12510/11 enters into a normal state of operation. The converter then operates continuously unless the voltage on VIN drops below 2.24 V or SHD is set low. UVLO hysteresis prevents the converter from dropping in and out of start-up, unintentionally locking up the system. Switching Node, 10 V/DIV Second Phase of Soft-Start First Phase of Soft-Start 100ms/DIV Choke Current, 100 mA/DIV SiP12511 SiP12511 VIN = 3.4 V; IOUT = 50 mA; VOUT = 24.4 V Figure 5. KEY APPLICATION CALCULATION INPUT CAPACITOR SELECTION The input bypass capacitor acts as an energy reservoir that satisfies the transient inductor current needs each time the switch turns on. In effect, the input capacitor is responsible for reducing the input voltage ripple and the amount of EMI that is inevitably passed to other circuitry on that line. For this purpose, a 1 µF capacitance minimum is recommended as input capacitor. A ceramic capacitor is recommended. If preferred, tantalum capacitors may be used instead of ceramics. OUTPUT CAPACITOR SELECTION To curb output voltage ripple, a multi-layer ceramic capacitor should be used as the output filter capacitor. Ceramic capacitors are favored for their low ESR (equivalent series resistance) and high resonance frequency, which makes them Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 www.vishay.com 5 New Product SiP12510/11 SiP12510/11 Vishay Siliconix ideal for high frequency switching converters. A high ESL (equivalent series inductance) can give rise to ringing in the low megahertz region and a high ESR could reduce phase margin and potentially cause instability of the design. In addition, the ripple current flowing through the capacitor's ESR causes power dissipation and heats up the capacitor internally. If the ripple current ratings of the capacitor are exceeded, the excessive temperature could shorten the expected life of the capacitor. If a high value capacitor is required for improved transient response, to keep component costs down and to save PC board real estate, tantalum capacitor may be used in parallel with ceramics. If the maximum tolerated ripple current (IP-P) and ripple voltage (VOUT) design specifications are known, the maximum tolerated ESR on the output capacitor and its value can be calculated using the following formulas: C OUT ( MIN ) = and I OUT ( MAX ) × D MAX fSW × VOUT ESR ( MAX ) = and VOUT I DIODE( MAX ) = 1 × I OUT ( MAX ) + I P- P 1 - D MAX 2 DUTY CYCLE CALCULATION In continuous mode of operation, the maximum duty cycle of a boost switching regulator determines the maximum amount of boost (VOUT/VIN) attainable and can be calculated using the expression VOUT + VDIODE - VIN ( MIN ) VOUT + VDIODE Where VDIODE is the forward bias voltage of the schottky diode and VIN(MIN) is the minimum operating input voltage of the converter. DMAX = 1 - Efficiency × VIN VOUT The above equation yields only an approximation of the duty cycle since it ignores power loss terms resulting from wire losses in the inductor, switching losses of the internal FET, and capacitor ripple current losses due to their inherent nonzero ESR. A more accurate estimate of the duty cycle can be www.vishay.com 6 DIODE SELECTION A schottky diode is recommended for use as the external rectifier. Schottky diodes are typically preferred in DC-DC conversion applications because of their low forward voltage drop and fast recovery time, which allows for high frequency switching. In choosing a diode, ensure that the diode's reverse breakdown voltage exceeds the intended VOUT of design and that its current rating is greater then the peak inductor current. SiP12510/11 SiP12510/11 have the typical 0.55 A maximum over current limiting function for the integrated power MOSFET. The maximum current of the inductor is limited to 0.55 A maximum typically. The relationship for the maximum diode current, maximum switch current and output current can be expressed as the following: I Q( MAX ) = I DIODE( MAX ) = I L ( MAX ) Where IOUT(MAX) is the maximum output current. D(MAX) is the maximum duty cycle. fSW is switching frequency. DVOUT is the output ripple voltage. COUT(MIN) is the minimum output capacitance. ESR(MAX) is the maximum equivalent resistance of the output capacitor. IP-P is the peak-to-peak value of the choke current. The formulas above are used to figure out the minimum out capacitance value. To reduce the out ripple voltage and improve the stability, it is recommended to used the larger capacitance for output capacitor. D(MAX) = determined byAnd by using the provided efficiency curves to approximate efficiency for a given input and output voltage. I OUT (1 - D MAX ) × DMAX × VOUT + 1 - DMAX 2 × L × fSW The average current rating of the output diode shall be equal to the output current. This is because the energy in the inductor only delivers to the output through the output diode when the switch is off. I DIODE(AVG) = I OUT For typical application of SiP12511 SiP12511, MBR0540 MBR0540 is recommended. For typical application of SiP12510 SiP12510, MBR0530 MBR0530 is recommended. INDUCTOR SELECTION An inductor is one of the energy storage components in a converter. Choosing an inductor means specifying its size, structure, material, inductance, saturation level, DC-resistance (DCR), and core loss. Choosing the right inductor is not a simple task and involves trade-offs in performance. The following are some key parameters that should be focused on. In PWM mode, inductance has a direct impact on the ripple current. The inductor value can be calculated as L= D × (VIN - VSW) I P-P × fSW Where VSW is the voltage drop across the switch in its onstate, fsw is the switching frequency, and D is the duty cycle. Higher inductance means lower ripple current, lower rms current, lower voltage ripple on both input and output, and higher efficiency, unless the resistive loss of the inductor dominates the overall conduction loss. However, higher inductance also means a larger inductor size and a slower transient response. For fixed line, load, and frequency conditions, higher inductance results in a lower peak current for each pulse and a higher load capability. The saturation current is another important parameter inchoosing inductors. Note that the saturation levels specified in data sheets are Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 New Product SiP12510/11 SiP12510/11 Vishay Siliconix maximum currents. For a DC-DC converter operating in PWM mode, it is the maximum peak inductor current that is relevant, and which can be calculated using these equations: I L ( MAX ) = or I OUT ( MAX ) I P - P + 1 - DMAX 2 I D ×V I L ( MAX ) = OUT ( MAX ) + MAX IN ( MIN ) 1 - DMAX 2 × fSW × L Where IL(MAX) is the maximum current in the choke. IOUT(MAX) is the maximum output current. This peak current varies with inductance tolerance and other errors, and the rated saturation level varies over temperature. So a sufficient design margin is required when choosing current ratings. A high-frequency core material, such as ferrite, should be chosen, the core loss could lead to serious efficiency penalties. The DCR should be kept as low as possible to reduce conduction losses. VIN D1 L1 10 µH C1 1 µF SHD VOUT U1 4 DS1 DS2 VOUT DS6 DS3 DS7 DS4 SiP12510 SiP12510 SiP12511 SiP12511 5 DS5 DS8 1 LX SHD R1 0 C2 1 µF 6 VIN 3 FB GND 2 0.1 V RFB = 5 Note: Populate R1 for SiP12510 SiP12510 only Figure 6. KEY APPLICATION CONSIDERATION LAYOUT CONSIDERATIONS In high frequency switching regulators such as the SiP12510/11 SiP12510/11, great attention must be given to the layout process in order to ensure stable operation and minimize noise. Since most power traces in step up converters carry pulsating current, energy stored in trace inductance during the pulse can cause high-frequency ringing with input and output capacitors. This effect can generally be curbed by minimizing the length and increasing the width of power traces. To minimize stray capacitance and even more importantly, parasitic trace inductance, all components must be kept as close to the switcher as possible. Of special importance, is the path between the switching node LX, D1, C1,C2, and ground of the regulator; the length of this path must be kept as small as possible since any parasitic inductance in series with the diode and output capacitance will increase noise and produce ringing in the circuit. Pulsating currents in the ground trace can cause voltage drops due to trace resistance and cause ground bounce. For this reason, it is strongly recommended to use a separate ground plane. As an example, Figure 6 and 7 demonstrate a recommended schematic and related layout of components. It is urged that this layout be followed closely as possible to obtain best performance. Figure 7. LED CURRENT CONTROL The SiP12510/11 SiP12510/11 is a white LED driver. The low feedback voltage of 0.1 V is designed to reduce losses outside of the white LEDs and thus improve overall circuit efficiency. The LED current is set by the small sense resistor on FB and can be calculated using the following expression: I LED = VREF 0.1 V = R FB R FB In order to have accurate LED current, use of 1 % precision resistor is recommended. As shown in Figures 1 and 2, the SiP12510 SiP12510 can be used to drive four LEDs in series or to drive parallel strings of LEDs. And SiP12511 SiP12511 can be used to drive up to seven LED in series. Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 www.vishay.com 7 New Product SiP12510/11 SiP12510/11 Vishay Siliconix WHITE LED BRIGHTNESS CONTROL Figures 8 and 9 delineate two possible brightness controlschemes. In Figure 8, a PWM signal is injected into the shutdown pin. The average LED current is proportional to the duty cycle of the PWM signal and thus, the brightness will vary from low to high as the duty cycle of the PWM signal is increased. The frequency of the PWM signal has to be low enough to allow the part to undergo soft-start and fully power up at each cycle. A frequency of 100 Hz to 500 Hz is, therefore, recommended. The magnitude of the PWM signal should be higher than the maximum enable voltage of SHD pin, in order to let the dimming control perform correctly. In Figure 9, a more analog approach to brightness control. As the control voltage VCTRL is increased from 0 V, the voltage drop across R2 and R3 increases driving voltage on node A low thereby reducing current through the White LEDs and dimming brightness. Reducing VCTRL to about 0 V, will turn the LEDs fully on with 20 mA of current. The equation for the LED current can be expressed as 0.1 V R 2 (0.1 V - VCTRL) + × 5 R3 5 I LED = Figure 10 demonstrates a more practical approach for dimming control,which is really the synthesis of the two ideas demonstrated above. In this approach, a filtered PWM signal acts as a DC voltage to control the brightness of the LEDs. It is recommended that PWM signal with frequency higher than 22 kHz be used. Figures 11 illustrate another ideal to power up WLED and additional load, which required constant output voltage. POWER DISSIPATION CONSIDERATIONS An important consideration when designing power converters is the maximum allowable power dissipation of a part. The maximum power dissipation in any application is dependant on the maximum junction temperature, TJ(MAX) = 125 °C, the junction-to-ambient thermal resistance for the TSOT-23 TSOT-23 package, J-A = 150 °C/W, and the ambient temperature, TA, which may be formulaically expressed as: P( MAX ) = TJ ( MAX ) - TA 125 °C - TA = J - A 150 °C /W It then follows that, assuming an ambient temperature of 70 °C, the maximum power dissipation will be limited to about 0.37 W. In the event that the power dissipation exceeds the value specified above and the die temperature reaches 165 °C, the internal thermal protection circuitry will ensure safe operation by turning off the internal MOSFET, thereby maintaining junction temperature at a safe level. In this state, only the system monitor circuitry will be active. Once the temperature of the SiP12510/11 SiP12510/11 drops below 145 °C, the SiP12510/11 SiP12510/11 re-enters soft-start mode and resumes normal operation. WHITE LED BRIGHTNESS CONTROL SCHEMATIC PWM ON/OFF CONTROL When low frequency PWM signal is available. MBR0540 MBR0540 L VIN 10 µH CIN 1 µF PWM FEEDBACK VOLTAGE CONTROL When high frequency PWM signal is available. VOUT COUT 1 µF 6 MBR0540 MBR0540 VIN 10 µH CIN 1 µF LX SHD 1 4 3 5 5 FB VOUT LX SHD 1 SiP12511 SiP12511 SiP12511 SiP12511 100 - 500 Hz COUT 1 µF 6 VIN VIN 4 VOUT L FB VOUT 3 0.1 V GND GND 2 2 R2 0.1 V 10 k RFB = 5 R3 20 k A RFB = 5 VCTRL Figure 8. www.vishay.com 8 Figure 9. Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 New Product SiP12510/11 SiP12510/11 Vishay Siliconix LINEAR FEEDBACK VOLTAGE CONTROL When PWM signal is not available. VIN MBR0540 MBR0540 L MBR0540 MBR0540 L VIN VOUT VOUT 10 µH 10 µH CIN 1 µF PWM CONTROL WITH CONSTANT OUTPUT VOLTAGE When the output is used to power up another load. CIN 1 µF COUT 1 µF 6 COUT 1 µF 6 VIN R1 VIN 4 4 1 LX SHD 5 VOUT 3 FB VOUT 1 SiP12511 SiP12511 SiP12511 SiP12511 5 LX SHD FB 3 0.1 V 0.1 V RLOAD RFB GND 2 GND 2 R2 R3 20 k A 10 k RFB = 5 PWM CDC RDC > 22 KHz 0.1 µF R3 PWM > 22 kHz Figure 10. Figure 11. TYPICAL CHARACTERISTICS 0.110 0.105 0.105 Feedback Voltage (V) Feedback Voltage (V) 0.110 0.100 0.100 0.095 0.095 0.090 - 40 0.090 2.5 3 3.5 4 4.5 5 5.5 6 - 20 0 20 40 60 80 100 VIN (V) Feedback vs. Input Voltage 120 Temperature (°C) Feedback Voltage vs. Temperature 95 0.12 94 92 0.08 VIN = 2.6 V R DS(on) () Maximum Duty Cycle (%) 0.10 93 91 90 VIN = 5 V 89 0.06 0.04 88 0.02 87 86 - 40 - 20 0 20 40 60 80 100 120 0.00 - 40 - 20 0 20 40 60 80 100 Temperature (°C) Maximum Duty Cycle vs. Temperature 120 Temperature (°C) Switching on Resistance vs. Temperature Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 www.vishay.com 9 New Product SiP12510/11 SiP12510/11 Vishay Siliconix TYPICAL CHARACTERISTICS 0.58 1400 0.56 Switch Current Limits (A) 0.60 1450 Switching Frequency (kHz) 1500 1350 1300 VIN = 2.6 V 1250 1200 VIN = 5 V 1150 0.54 0.52 0.50 0.48 0.46 1100 0.44 1050 0.42 1000 - 40 - 20 0 20 40 60 80 100 0.40 - 40 120 - 20 0 20 40 60 80 100 120 Temperature (°C) Temperature (°C) Switch Current Limit vs. Temperature Switching Frequency vs. Temperature 0.685 Switching Current Limit (A) 0.635 0.585 0.535 0.485 0.435 0.385 2.5 3 3.5 4 4.5 VIN (V) 5 5.5 6 Switch Current Limit vs. Input Voltage www.vishay.com 10 Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 New Product SiP12510/11 SiP12510/11 Vishay Siliconix TYPICAL WAVEFORMS 100 100 VIN = 6 V VIN = 6 V 90 90 VIN = 3.6 V 80 VIN = 4.2 V VIN = 3.6 V Efficiency (%) Efficiency (%) 80 70 60 VIN = 4.2 V 70 60 50 50 40 40 SiP12511 SiP12511: 8 LEDs SiP12511 SiP12511: 7 LEDs 30 30 0 5 10 15 20 I OUT m(A) 25 30 0 5 Efficiency vs. Output Current 10 15 20 I OUT m(A) 30 Efficiency vs. Output Current 100 100 VIN = 6 V VIN = 6 V 90 90 80 80 VIN = 4.2 V VIN = 3.6 V 70 60 VIN = 4.2 V VIN = 3.6 V Efficiency (%) Efficiency (%) 25 70 60 50 50 40 40 SiP12511 SiP12511: 6 LEDs SiP12510 SiP12510: 4 LEDs 30 30 0 5 10 15 20 25 0 30 5 10 15 20 25 30 I OUT m(A) I OUT m(A) Efficiency vs. Output Current Efficiency vs. Output Current 100 VIN = 4.2 V 90 Efficiency (%) 80 VIN = 6 V 70 VIN = 3.6 V 60 50 40 SiP12510 SiP12510: 3 LEDs 30 0 5 10 15 I OUT m(A) 20 25 30 Efficiency vs. Output Current Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 www.vishay.com 11 New Product SiP12510/11 SiP12510/11 Vishay Siliconix TYPICAL WAVEFORMS VIN 2 V/DIV Switching Node, 20 V/DIV SHD 2 V/DIV Output Voltage (AC Coupling) 50 mV/DIV Switching Node, 10 V/DIV Choke Current 200 mA/DIV Choke Current 200 mA/DIV 2 µs/DIV 500 ns/DIV Steady State Operation VIN = 2.5 V, 7 LEDs In Series. IOUT = 20 mA Power-Down VIN = 2.5 V, 7 LEDs In Series. IOUT = 20 mA VIN 5 V/DIV VIN 5 V/DIV Switching Node, 20 V/DIV Switching Node, 20 V/DIV Output Voltage (AC Coupling) 50 mV/DIV Output Voltage (AC Coupling) 200 mV/DIV Choke Current 200 mA/DIV Choke Current 500 mA/DIV 1 µs/DIV 500 ns/DIV Steady State Operation VIN = 6 V, 7 LEDs In Series. IOUT = 20 mA Steady State Operation VIN = 6 V, 7 LEDs In Series, 5 Series In parallel, IOUT = 100 mA VIN 2 V/DIV Switching Node, 20 V/DIV SHD 2 V/DIV Switching Node, 10 V/DIV Output Voltage (AC Coupling), 20 mV/DIV Choke Current 200 mA/DIV Choke Current, 200 mA/DIV 50 µs/DIV 1 µs/DIV Steady State Operation VIN = 2.5 V, 7 LEDs In Series, 5 Series In parallel, IOUT = 100 mA Start-Up VIN = 6 V, 7 LEDs In Series. IOUT = 20 mA SHD 2 V/DIV SHD 2 V/DIV Switching Node, 10 V/DIV Switching Node, 10 V/DIV Choke Current 200 mA/DIV 50 µs/DIV Start-Up VIN = 2.5 V, 7 LEDs In Series. IOUT = 20 mA www.vishay.com 12 Choke Current 200 mA/DIV 2 µs/DIV Power-Down VIN = 6 V, 7 LEDs In Series. IOUT = 20 mA Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 New Product SiP12510/11 SiP12510/11 Vishay Siliconix TYPICAL WAVEFORMS VIN 5 V/DIV VIN 5 V/DIV Switching Node, 20 V/DIV Switching Node, 20 V/DIV Output (AC Coupling), 50 mV/DIV Output (AC Coupling), 200 mV/DIV Choke Current 500 mA/DIV Choke Current, 200 mA/DIV 500 ns/DIV 500 ns/DIV Steady State Operation VIN = 6 V, 8 LEDs In Series, IOUT = 20 mA Steady State Operation VIN = 6 V, 8 LEDs In Series, 3 Series in parallel, IOUT = 60 mA SHD 5 V/DIV SHD 5 V/DIV Switching Node, 20 V/DIV Switching Node, 20 V/DIV Choke Current, 200 mA/DIV Choke Current 200 mA/DIV 50 µs/DIV 2 µs/DIV Start-Up VIN = 6 V, 8 LEDs In Series, 3 Series in parallel, IOUT = 60 mA Power-Down VIN = 6 V, 8 LEDs In Series, 3 Series In parallel, IOUT = 60 mA DESIGN CHECKLIST AND REVIEW To design any power conversion circuit, it is necessary to verify the proper functionality on the bench to ensure the solidity of the design. The following are the some tests recommend proving the proper functionality of SiP12510/11 SiP12510/11 design. 1. Stability: Power up the SiP12510/11 SiP12510/11 white LED driver design. Monitor the switching node waveform in LX pin. Use the current probe to monitor the current waveform in the choke. Change the input voltage and white LEDs load in the designed circuit. Make sure the waveforms are stable in any designed line/load condition. Instable waveforms sometimes are caused by the improper design of the choke current. The current limiting circuit tries to shut down itself. 2. Current limit threshold: Make sure the peak current of the choke does not exceed the minimum current limit of the builtin switch in any designed line/load conditions. It is recommended to have some design margin. Exceed of the current limit of the switch will result instable switching waveforms. 3. Duty Cycle: Make sure the duty cycle is not locked in any designed line/load condition. The operating duty cycle in any designed line/load condition must be less than 85 %, which is the minimum value of the maximum duty cycle. 4. Output Voltage Check: Monitor the output voltage of the converter. Make sure the output voltage is less than the minimum over voltage protection threshold voltage, which is 17 V for SiP12510 SiP12510 and 27.5 V for SiP12511 SiP12511. Vishay Siliconix maintains worldwide manufacturing capability. Products may be manufactured at one of several qualified locations. Reliability data for Silicon Technology and Package Reliability represent a composite of all qualified locations. For related documents such as package/tape drawings, part marking, and reliability data, see http://www.vishay.com/ppg?74498. Document Number: 74498 S-70770 S-70770­Rev. A, 30-Apr-07 www.vishay.com 13 Legal Disclaimer Notice Vishay Notice Specifications of the products displayed herein are subject to change without notice. Vishay Intertechnology, Inc., or anyone on its behalf, assumes no responsibility or liability for any errors or inaccuracies. Information contained herein is intended to provide a product description only. No license, express or implied, by estoppel or otherwise, to any intellectual property rights is granted by this document. Except as provided in Vishay's terms and conditions of sale for such products, Vishay assumes no liability whatsoever, and disclaims any express or implied warranty, relating to sale and/or use of Vishay products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright, or other intellectual property right. The products shown herein are not designed for use in medical, life-saving, or life-sustaining applications. Customers using or selling these products for use in such applications do so at their own risk and agree to fully indemnify Vishay for any damages resulting from such improper use or sale. Document Number: 91000 Revision: 08-Apr-05 www.vishay.com 1