APT9403 ARF440 ARF441 EL7144C ARF440/ARF441 C4-C27 SKR330M1HE11V TT250M20A - Datasheet Archive

 

By: Kenneth Dierberger Simple and Inexpensive High-Efficiency Power Amplifier Using New APT MOSFETs Presented at RF Expo East

 

APT9403 APT9403 By: Kenneth Dierberger Simple and Inexpensive High-Efficiency Power Amplifier Using New APT MOSFETs Presented at RF Expo East November 1994 Simple and Inexpensive High-Efficiency Power Amplifier Using New APT MOSFETs Kenneth Dierberger Applications Engineering Manager Advanced Power Technology Inc. 405 SW Columbia St. Bend, Oregon 97702 USA Frederick H. Raab, Ph.D. Green Mountain Radio Research Co. 50 Vermont Ave., Fort Ethan Allan Colchester, Vermont 05446 Lee Max Independent Consultant 6284 Squiredell Dr. San Jose, California 95129 ABSTRACT 2. BASIC DESIGN CONSIDERATIONS This note describes a simple and inexpensive highefficiency RF power amplifier based upon the new Advanced Power Technology (APT) RF-power MOSFETs. Its key features are a transformerless driver, low-cost RF-power MOSFETs, a simple output transformer, and Class-D operation. The PA and driver operate from +12 and +50V and require an RF input of only 10mW. They produce an output of 250W or more at frequencies from 1.8 to 13.56MHz. Class-D power amplifiers employ two transistors in a push-pull configuration. The transistors are driven to act as switches and generate a square-wave voltage. The fundamental-frequency component of the square wave is passed to the load through a filter. Power output is controlled by varying the supply voltage. A class-D PA is ideally 100-percent efficient at all amplitudes and in spite of load reactance. In practice, it is significantly more efficient than a similar class-B PA, especially for lower amplitudes and reactive loads. 1. INTRODUCTION Drain-Load Line and Turns Ratio This note describes a simple and inexpensive highefficiency RF power amplifier based upon the new APT RF-power MOSFETs. Its key features are: · A transformerless driver, · Low-cost RF-power MOSFETs, · Simple output transformer, and · Class-D operation. The PA and driver operate from +12 and +50V and require an RF input of only 10mW. They produce an output of 250W or more at frequencies from 1.8 to 13.56MHz. The basic concept ensures an amplifier that is inherently inexpensive and easy to manufacture. In the prototype described here, however, no attempt has been made to minimize the number of components and every attempt has been made to ensure good RF grounds and bypasses. The power output of a class-D PA [1] [2] is: PO = 8 v eff 2 p 2 2R (1) where the effective supply voltage (for MOSFETs) is: V eff = V DD R R + R ON (2) Above, R is the drain-load line (seen by one drain with the other open) and RON is the on-state drain-source resistance. For the ARF440 ARF440 and ARF441 ARF441, R ON»0.8W. To obtain a 250W output with a 50V supply thus requires R£6.4W. For a simple transformer, the drain-load line is related to the load Ro by: ( ) 2 R = m n Ro where m and n are the numbers of turns in the primary and secondary windings, respectively. For an RF transformer, m and n must be small integers. For a 50W load, a bit of iteration yields m =1, n =3, and R=5.56W. Ignoring transformer, switching, and capacitance losses, the efficiency of the PA is: h= V eff = 0.874 V DD 3. CIRCUIT (3) The circuit of the power amplifier and driver is shown in Figure 1; parts are identified in Tables 1 and 2. All stages are ac-coupled. Consequently, the unexpected absence of an input signal results only in an inactive circuit with minimal current flow and power consumption. The total cost of all of the parts in the prototype breadboard is about $190, based upon quantities ranging from 1 to 100. (4) Pre driver The effects of switching and drain capacitance can be predicted by the equations given in [2]. These calculations yield a 76.2 percent efficiency for operation at 13.56MHz, excluding losses in the transformer and output filter. With typical losses in those components, an efficiency of 70 percent is expected. The Elantec EL7144C EL7144C is intended for use as a gate driver. The internal Schmitt trigger allows it to serve as hard limiter, and the presence of both inverting and non-inverting inputs allows a pair to serve as a phase splitter. Gate Voltage and Current The peak drain current [1] [2] is: i Dmax = 4 v eff = 10.01A p R The pre driver uses a pair of low-cost ICs (U1 and U2 in Figure 1) rather than the conventional RF transformer to provide out-of-phase driving signals for the two final MOSFETs. It also provides hard limiting (sinewave to square-wave conversion) of the input signal. (5) which corresponds to Idc =6.37A. The data sheets for the ARF440/ARF441 ARF440/ARF441 show that a gate-source voltage of 9 to 10V should be sufficient for minimum RON at iDmax=10A. Since the threshold voltage is about 3.5V, the RF drive voltage must be about 6.5V or slightly more. The data sheets give typical small-signal active-region gate-source and gate-drain capacitances of 700 and 55pF, respectively. The input capacitance is, however, the sum of small-signal capacitances only while the MOSFET is in the cut-off region. As the MOSFET enters the active (velocity-saturation) region, the gatedrain capacitance is Miller-multiplied by the voltage gain (plus one). Finally, as the MOSFET enters the linear region (resistive region), the gate-drain capacitance inflates by a factor of five or so. As a result, the effective gate-drain capacitance is close to 2600pF during the switching process [3]. The total gate charge is 37nC at 10V. Transitioning the gate voltage in one tenth of the RF period (73.7ns at 13.56MHz) thus requires an average of 5A of gate current. A low driving resistance and low inductance are clearly required. The RF input is ac-coupled to the non-inverting input of U1 and the inverting input of U2. Adjustment of the biases via R6 and R8 allows the transition points to be selected to produce the desired duty ratio. For operation at the lower frequencies (£4MHz), the duty ratio can be set to 50:50. In this case, the phase error between the two EL7144s is about 0.5ns. At higher frequencies (³10MHz), the difference in the turn-on and turn-off delay times of the ARF440 ARF440 and ARF441 ARF441 can cause both to be on at the same time (Appendix A). The associated increase in current consumption is eliminated by adjusting the duty ratios of U1 and U2 so that the final MOSFETs operate with 50:50 duty ratios. This requires adjusting R6 and R8 so that the low-state pulse width is about 12ns shorter than the high-state pulse width. Some phasing error (up to 25°) is unfortunately introduced by this adjustment. The EL7144s have high input impedances, so R5 provides a 50W input impedance for the signal source. Input signals in the range of 10 to 100mW are satisfactory, allowing this PA to be driven directly from a laboratory signal generator. The best switching speed is obtained with V DD1=12V. J3 J4 (VDD1 ) J5 J6 (VDD2 ) J7 J8 (VDD3 ) D1 C3 R1 C C1 D2 C2 R3 D3 R2 L1 A R6 C D B B R10 E C R12 C14 C7 C C15 B A R7 C5 E R18 R4 C24 C16 C6 R11 R19 Q1 1 8 2 U1 7 3 6 4 5 C4 J1 RF INPUT Q5 C12 R13 R5 C22 Q2 C13 C26 T1 J2 RF OUTPUT L2 C28 B A C9 C27 C10 C17 C8 C21 1 8 2 U2 7 3 6 4 5 Q3 C23 Q6 R15 R9 Q4 R21 C18 C19 C11 R17 C20 C25 D R8 B R14 E D R16 D R20 Figure 1. Circuit of power Amplifier and driver REFERENCE DESIGNATOR PART DESCRIPTION C1,C2 C3 C4-C27 C4-C27 C28 33mF, 50V electrolytic, Mallory SKR330M1HE11V SKR330M1HE11V 20mF, 250V electrolytic, Mallory TT250M20A TT250M20A 0.1mF, 50WV chip, ATC 200B104NP50X 200B104NP50X See Table 2. D1,D2,D3 5.1V, 0.25W Zener, 1N751A 1N751A J1,J2 BNC female bulkhead Recommended: Amphenol 31-5538 Used: RF Industries RFB-1116S/UG RFB-1116S/UG European-style binding post, Johnson 111-0104 J3-J8 L1 L2 3.5mH, 7 turns #24 enameled wire on Ferroxcube 768XT188 768XT188 4C4 toroid See Table 2. Q1,Q3 Q2,Q4 Q5 Q6 p-channel MOSFET, Siliconix 2N7016 2N7016 n-channel MOSFET, Siliconix 2N7012 2N7012 n-channel MOSFET, APT ARF440 ARF440 n-channel MOSFET, APT ARF441 ARF441 R1,R2 R3 R4 R5 R6,R8,R10,R12,R14,R16,R18,R20 R7,R9,R11,R13,R15,R17,R19,R21 330W RC07 220W RC07 10KW RC07 51W RC07 1KW trimpot, Bournes 3299X-1-102 3299X-1-102 4.7kW RC07 T1 One Ceramic Magnetics 3000-4-CMD5005 3000-4-CMD5005 block of CMD5005 CMD5005 ferrite wired per Figure 3 U1,U2 Schmitt trigger/limiter, Elantec EL7144C EL7144C Heat sink for DIP IC Heat sink for Q5 and Q6 Aavid 5802 clip-on Aavid 61475, 6.5-in wide by 4-in long IC socket, 8-pin DIP (4) Augat 208-AG190C 208-AG190C PC board Approximately 6.5-in wide by 8-in long Plastic bracket for L2 Plastic bracket for C28 Cut from plastic L Cut from plastic L Feet (6) Aluminum threaded spacer, 4-40 x 2 in (Keystone 2205) Table 1. Components other than tuning. 1.8MHz 180pF, 2.5-kV padder, FW Capacitors AP051HV AP051HV L2: 5.7mH, Micrometals T200-2 T200-2 toroid (2-in O.D.) with 20 turns of 24-AWG 24-AWG enameled wire 90pF, 2.5kV padder, FW Capacitors AP031HV AP031HV L2: 2.93mH, Micrometals T200-2 T200-2 toroid (2-in O.D.) with 14 turns of 20-AWG 20-AWG enameled wire 86pF, 2.5kV padder, FW Capacitors AP031HV AP031HV L2: 2.93mH, Micrometals T200-2 T200-2 toroid (2-in O.D.) with 14 turns of 20-AWG 20-AWG enameled wire C28: 60pF, 2.5kV padder, FW Capacitors AP031HV AP031HV L2: 2.93mH, Micrometals T200-2 T200-2 toroid (2-in O.D.) with 14 turns of 20-AWG 20-AWG enameled wire C28: 13.56MHz 11.4mH, Micrometals T200-6 T200-6 toroid (2-in O.D.) with 32 turns of 24-AWG 24-AWG enameled wire C28: 12MHz L2: C28: 10MHz 354pF, 2.5kV padder, FW Capacitors AP091HV AP091HV C28: 7MHz 22mH, Micrometals T200-6 T200-6 toroid (2-in O.D.) with 52 turns of 24-AWG 24-AWG enameled wire C28: 3.5MHz L2: 47pF, 2.5kV padder, FW Capacitors AP031HV AP031HV Table 2. Tuning components. Driver The EL7144Cs provide sufficient drive for the ARF440 ARF440 and ARF441 ARF441 at frequencies up to 4MHz. Operation at higher frequencies requires a driver with a higher current rating and lower resistance. The driver consists of two complementary pairs (Q1-Q2 and Q3-Q4 in Figure 1). The prototype uses Siliconix 2N7016 2N7016 and 2N7012 2N7012, which are in quarter-size DIP packages. A variety of similar complementary MOSFETs can also be used (e.g., IRFD110 IRFD110, IRFD9120 IRFD9120). The prototype provides adjustable bias for each MOSFET. The objective of the bias voltage is to place the gate voltage just below the threshold. Thus R12 is adjusted to set the gate of Q2 at 3V, while R10 is adjusted to set the gate of Q1 at V DD2-3V. Adjustment is not critical and these potentiometers can be replaced by fixed dividers. currents are individually adjustable via R18 and R20 and should be set to about 0.1A each (i.e., just on the verge of conduction). This results in gate bias of about 3.5 to 3.8V. The variable gate biases provide a convenient means of checking final MOSFETs Q5 and Q6 during development. The ac coupling also provides some isolation of the driver MOSFETs in case of a shortcircuit failure of the final MOSFETs. It is, however, possible to simplify the circuit by direct-coupling the output of the driver to the gate of the final. If this is done, the bias on Q2 and Q4 should be increased slightly to ensure that Q5 and Q6 are cut-off in the absence of an input signal. Output transformer T1 is constructed by winding #22 insulated wire through one block of CMD5005 CMD5005 Q5 GROUND A supply voltage of V DD2=12V provides sufficient drive for the final amplifier. This voltage is conveniently the same as VDD1. VDD3 FILTER Final Amplifier Q6 The final amplifier is based upon the ARF440 ARF440 and ARF441 ARF441 symmetric-pair MOSFETs. The quiescent Figure 2. Construction of Output Transformer T1. ferrite as shown in Figure 2. The use of different colors facilitates identification of the leads. The VSWR and overrating factor are less than 1.9 and 1.5, respectively, from 1 to 14MHz. Capacitors C26 and C27 ensure a good RF ground at the center of the primary winding. Inductor L1 keeps RF out of the power-supply line; its value is not critical. The prototype uses simple series-tuned circuits (Table 2) with Q =5 at the frequency of operation (except that the 13.56MHz inductor is used at 10 and 12MHz). Tuning is accomplished by adjusting padder C28 for maximum output power or maximum dc-input current. Other output filters or matching networks can be used provided they include a series inductor on the transformer side to prevent current from flowing at the harmonic frequencies. 4. LAYOUT The general layout of the principal components is shown in Figure 3. MOSFETs Q5 and Q6 are separated by about 0.8 inchs so that their drain leads are aligned with the leads from T1. The drivers and pre-drivers are installed roughly in line with the leads from Q5 and Q6. The prototype breadboard is constructed on a 6.5 inchs by 8 inchs piece of PC board. All wiring is done on or above the surface to simulate a PC board with traces on one side and a continuous ground plane on the other. For convenience in experimentation, U1, U2, Q1, Q2, Q3, and Q4 are socketed. Socketing is not, however, recommended for a final layout as it adds inductance between active devices. The input to U1 and U2 is high-impedance and therefore not especially critical. Chip capacitors C5, C6, C9, and C10 are installed as close as possible to the power-supply leads. A clip-on heat sink is placed over each IC to provide adequate heat dissipation. The pre driver outputs are connected to the driver inputs through short, wide (low-inductance) traces. Chip capacitors C16 and C21 are again installed as close as possible to the drain leads from Q1 and Q3. The outputs of the drivers are connected to the gates of Q5 and Q6 through short, wide traces. A clip-on heat sink is placed over each pair to provide adequate heat dissipation. The heat sink is mounted flush with the bottom of the PC board. Holes are cut for Q5 and Q6, but the integrity of the ground plane is otherwise not interrupted. Connection pads are cut near the gates and drains. The leads of Q5 and Q6 are trimmed at the point they narrow and then bent downward slightly to contact the PC board. Output transformer T1 lies flat on the PC board and is held in place adequately by its leads. Tuning elements L2 and C28 are supported on plastic L brackets. 5. PERFORMANCE Tuned Output R6 R10 R12 R16 A tuned output is used for transmitters and Q5 U1 Q1 Q2 U2 Q3 Q4 C26,C27 J1 T1 L1 R8 R14 R16 R20 Q6 Figure 3. General Layout of Amplifier (not to scale) C28 J2 resonant loads. The series-tuned output reduces the levels of the harmonics so that they contribute negligibly to the output power, which is measured by a Bird 4421 wattmeter. The waveforms at 1.8, 7, and 13.56MHz are shown in Figures 4, 5, and 6. The waveforms include the predriver output vEL , final gate v GS , final drain v D S, transformer output v S and filter output v O . All waveforms except vS are obtained with V DD=40V; vS is obtained with V D D =20V to avoid clipping by the oscilloscope. Turn-on begins shortly after the gate voltage begins to rise and flattens the driving waveform because of the Figure 4. Waveform for Tuned Load at 1.8MHz Miller effect and inflation of the gate-drain capacitance. The transients in vEL and v GS are probably due to the length of wire connecting the driver and gate, and/or the voltage developed across the internal inductances of the MOSFETs. The transients in the drain voltage are the normal consequence of the MOSFET switching faster than the rise time of the transformer. The transient frequency of 56MHz corresponds roughly to the resonance of the 155pF typical drain capacitance and the measured 40nH of transformer inductance. The peak voltage remains well within the ratings of the ARF440 ARF440 and ARF441 ARF441. The transient is prevented from reaching the load by the output filter. Figure 5. Waveform for Tuned Load at 7.0MHz be reduced slightly (Table 3) to maintain safe drain currents (iDmax£11A, Idc£7A). The output power drops accordingly. The most probable explanation of the lower-than-expected efficiency at higher frequencies are imperfect timing of the drive and the long tail in the turn-on of the MOSFET. The variations of efficiency and output voltage with supply voltage are shown in Table 4 and Figure 8. In contrast to class-A and -B PAs, the efficiency remains relatively constant and high for all output levels. The highest efficiency generally occurs at mid-range supply f, MHz V DD, V IDC, A Pi , W P o, W h 1.8 3.5 7.0 10.0 12.0 13.56 6.55 6.83 6.85 6.74 6.72 6.72 327.5 341.5 333.0 303.5 268.6 302.6 273.0 274.0 250.0 206.3 184.8 184.7 0.833 0.802 0.751 0.679 0.688 0.610 50.0 50.0 48.6 45.0 40.0 45.0 Table 3. Power Output and Efficiency for Tuned Load. Figure 6. Waveform for Tuned Load at 13.56MHz The variations of efficiency and power output with frequency are shown in Table 3 and Figure 7. Also included are predictions of performance for V DD=50V. These are based upon on resistance, switching time, and drain capacitance, but do not include losses from the transformer and output filter. The measured efficiency is based upon measured RF output and dc input to the final amplifier (Q5 and Q6) and include all losses. At the lower frequencies, V DD =50V. The power output and efficiency are in excellent agreement with the predictions, especially with a few percent of loss in the transformer and filter added. At higher frequencies, the efficiency is a little less than expected and V DD must Figure 7. Efficiency and Output versus Frequency for Tuned Load V DD,V 1.8 MHz vom, V h 7.0 MHz vom, V h 13.56 MHz vom, V h 10 20 30 40 45 47.8 48.6 50 34.4 68.6 102.4 135.0 149.7 158.1 0.823 0.845 0.853 0.847 0.837 0.833 34.6 69.6 102.3 133.4 148.0 0.705 0.751 0.745 0.755 0.753 36.7 69.2 101.1 125.4 135.9 158.1 0.751 165.2 0.833 0.573 0.613 0.645 0.626 0.610 Table 4. Output Voltage and Efficiency for Tuned Load. Untuned operation is used to deliver the maximum raw RF power to a resistive load. Tuning components L2 and C28 are omitted and the transformer output connected directly to the load. The measured output power includes all harmonics and is obtained from the rms voltage measured by an HP 54503 oscilloscope. Because both the drain voltage and the drain current are square waves, this mode of operation can provide 27 percent more RF power than tuned class D. In an ideal square wave, 81 percent of the power is at the fundamental frequency and 19 percent is in the harmonics. The peak drain current and dc-input current are (ideally) equal. The waveforms for 1.8, 7, and 13.56MHz are shown in Figures 9, 10, and 11. All of these waveforms are obtained with V DD=40V. Transients are still present in the drain waveform, but are greatly reduced in the output waveform because of the resistive loading (in contrast to the high impedance for tuned operation). Figure 8. Efficiency and Output versus Supply Voltage for Tuned Load voltages because the drain capacitance is reduced but the current is not yet high enough to increase Ron. The linearity for control and modulation is generally excellent. Untuned Output Figure 9. Waveforms for Untuned Load at 1.8MHz Figure 10. Waveforms for Untuned Load at 7.0MHz. The efficiency and power output are shown as functions of frequency in Table 5 and Figure 12. At the lower frequencies, V DD =50V produces outputs over 300W with efficiencies of 85 percent. At the higher frequencies, VDD is increased to maintain an output of 250W. Slightly higher efficiencies are generally obtained for outputs of about 100W. Driver The power consumption of the pre driver and driver are given in Table 6. The power dissipations are slightly above those for static air and no heat-sinks. However, the use of the clip-on heat sinks and some moving air has afforded reliable operation. The driver current is roughly twice that required to charge the gate capacitance, so some shoot-through current (both MOSFETs on simultaneously) is no doubt present. This can probably be reduced by lowering the gate biases. Figure 11. Waveforms for Untuned Load at 13.56MHz. 6. COMMUNICATION APPLICATIONS The power amplifier and driver can be keyed for CW transmission by interruption of either the RF drive or the dc supply. Amplitude modulation is very linear (Figure 8) and can be accomplished efficiently by a class-S modulator [4], [5]. Production of single sideband and other complex signals can be accomplished by the Kahn envelope-elimination-andrestoration (EER) technique [6] [7]. 7. RECOMMENDATIONS The proof-of-concept prototype described here demonstrates that this simple and inexpensive PA can produce significant power with good efficiency at frequencies up to 13.56MHz. f, MHz V DD, V IDC, A Pi , W P o, W h 1.8 3.5 7.0 10.0 12.0 13.56 7.50 7.00 6.13 5.94 5.77 6.31 375.2 350.0 315.4 326.7 329.2 363.2 323.6 297.1 252.1 249.5 250.4 245.5 0.862 0.849 0.799 0.764 0.761 0.676 50.0 50.0 51.5 55.0 52.1 61.9 The following additional efforts are therefore recommended: · Add monostable (timer) circuits to the predriver so that both the phase and the width of the driving pulses can be controlled. · Identify and test suitable complementary pairs for the driver. · Prepare a new PC board with broad, lowinductance traces connecting the driver output and ARF440/ARF441 ARF440/ARF441 gates. Use surface mounting rather than sockets. Table 5 Power Output and Efficiency for Untuned Load. The ARF440 ARF440 and ARF441 ARF441 have ratings sufficient to support an output of 500W or more. To obtain this will require the improved driver and a different load line, as well as some other changes in components. The following additional efforts are therefore recommended: · Implement a new transformer for a 12.5W load line and compare its performance to that of a transmission-line transformer. · Using the selected 12.5W transformer, test operation at outputs up to 500W with supply voltage up to 100V. The switching times of the ARF440/ARF441 ARF440/ARF441 suggest that full-power operation (in class B) is possible to frequencies as high as 30MHz. This will require the transmission-line transformer mentioned above. The following efforts are recommended. · Implement a conventional drive circuit using a transmission-line transformer and gate swamping. · Test the PA in both class-D and class-B operation (similar to [2]). 8. REFERENCES Figure 12. Efficiency and Output versus Frequency for Untuned Load f, MHz 1.8 3.5 7.0 10.0 12.0 13.56 Idc1, A 0.11 0.16 0.27 0.32 0.40 0.45 P i1, W 1.32 1.92 3.24 3.84 4.80 5.40 Idc2, A 0.18 0.32 0.62 0.82 1.10 1.25 P i2, W 2.16 3.84 7.44 9.84 13.2 15.00 Table 6. Power Consumption of Drivers. It should nonetheless be possible to improve both efficiency and power output at the higher frequencies. [1] H. L. Krauss, C. W. Bostian, and F. H. Raab, Solid State Radio Engineering . New York: Wiley, 1980. [TP76-4 TP76-4] [2] F. H. Raab and D. J. Rupp, "HF power amplifier operates in both class B and class D," Proc. RF Expo West '93, San Jose, CA, pp. 114 - 124, March 17 - 19, 1993. [TP92-7 TP92-7] [3] K. Dierberger, "Gate drive design for large die MOSFETs," Application Note APT9302 APT9302, Advanced Power Technology, Bend, OR, 1993. Also Proc. PCIM'93 Europe . [H-1274 H-1274] [4] F. H. Raab and D. J. Rupp, "Class-S high-efficiency amplitude modulator," RF Design, vol. 17, no. 5, pp. 70 - 74, May 1994. [TP93-1 TP93-1] [5] F. H. Raab and D. J. Rupp, "High-efficiency amplitude modulator," Proc. RF Expo East '94, Orlando, FL, pp. ? - ?, Nov. 15 - 17 1994. [TP94-7 TP94-7] [6] F. H. Raab and D. J. Rupp, "High-efficiency singlesideband HF/VHF transmitter based upon envelope elimination and restoration," Proc. Sixth Int. Conf. HF Radio Systems and Techniques (HF'94) (IEE CP 392), York, UK, pp. 21 - 25, July 4 - 7, 1994. [TP94-1 TP94-1] [7] F. H. Raab and D. J. Rupp, "High-efficiency multimode HF/VHF transmitter for communication and jamming," Proc. MILCOM'94 , Ft. Monmouth, NJ, pp. 880 - 884, Oct. 2 - 5, 1994. [TP94-3 TP94-3] APPENDIX A. SWITCHING CHARACTERISTICS OF ARF441 ARF441 The basic switching characteristics of the ARF441 ARF441 and the complementary driver are shown in Figure 13. For this test, the driver and pre driver are connected as shown in Figure 1. However, the drain of Q6 is connected to VDD3 through a 12.5W RF load resistor. Figure 13. ARF441 ARF441 Switching Characteristics. The turn-on time is on the order of 6 to 7ns and suggests that full-power operation in class B is possible to frequencies as high as 30MHz. However, turn-off is delayed by about 11ns longer than turn-on. The MOSFET is biased at its threshold voltage, so the turn-on and turn-off transitions involve the same voltage change. However, the gate-drain capacitance is considerably smaller during turn-on than turn-off, resulting in a shorter transition. Transients may also contribute to this difference. The difference in turn-on and turn-off delays has important implications for operation of the amplifier at frequencies above about 7MHz. Drive signals with 50:50 duty ratios cause Q5 and Q6 to be on simultaneously during a portion of the RF cycle. The resultant build-up of current in T1 produces large transients and inefficiency. This problem is avoided by adjusting the predriver to produce shortened drive pulses so that the final MOSFETs operate with 50:50 duty ratios. The long tail in the turn-on characteristic appears to be associated with the slower increase in gate voltage as the MOSFET nears the linear region and the gatedrain capacitance inflates. It results in a higher effective on resistance for a period of time, hence lower efficiency at higher frequencies. (This effect is called "RF-saturation voltage" in BJTs.) The long tail can probably be made shorter by a better driver. 405 S.W. Columbia Street Bend, Oregon 97702 USA Phone: (541) 382-8028 Fax: (541) 330-0694 http://www.advancedpower.com Parc Cadera Nord - Av. Kennedy BAT B4 33700 Merignac, France Phone: 33-557 92 15 15 Fax: 33-556 47 97 61 Printed - March 1995